 |
|
| If you can't view the Datasheet, Please click here to try to view without PDF Reader . |
|
|
|
| If you can't view the Datasheet, Please click here to try to view without PDF Reader . |
|
|
| Datasheet File OCR Text: |
| a FEATURES Monolithic 16-Bit, Oversampled A/D Converter 8 Oversampling Mode, 20 MSPS Clock 2.5 MHz Output Word Rate 1.01 MHz Signal Passband w/0.004 dB Ripple Signal-to-Noise Ratio: 88.5 dB Total Harmonic Distortion: -96 dB Spurious Free Dynamic Range: 100 dB Input Referred Noise: 0.6 LSB Selectable Oversampling Ratio: 1 , 2 , 4 , 8 Selectable Power Dissipation: 150 mW to 585 mW 85 dB Stopband Attenuation 0.004 dB Passband Ripple Linear Phase Single +5 V Analog Supply, +5 V/+3 V Digital Supply Synchronize Capability for Parallel ADC Interface Twos-Complement Output Data 44-Lead MQFP High-Speed Oversampling CMOS ADC with 16-Bit Resolution at a 2.5 MHz Output Word Rate AD9260 FUNCTIONAL BLOCK DIAGRAM DRVDD AVDD AVDD AVDD AVSS AVSS AVSS DRVSS RESET/ SYNC DVSS DVDD VINA MULTIBIT SIGMA-DELTA MODULATOR DIGITAL DEMODULATOR 12-BIT: 20MHz STAGE 1:2X DECIMATION FILTER OTR VINB OUTPUT MODE MULTIPLEXER 16-BIT: 10MHz AD9260 16-BIT: 5MHz REFERENCE BUFFER STAGE 3:2X DECIMATION FILTER STAGE 2:2X DECIMATION FILTER OUTPUT REGISTER BIT1-BIT16 REF TOP REF BOTTOM COMMON MODE 16-BIT: 2.5MHz VREF SENSE REFCOM BANDGAP REFERENCE BIAS CIRCUIT CLOCK BUFFER MODE REGISTER DAV READ BIAS ADJUST CLK MODE CS PRODUCT DESCRIPTION The AD9260 is a 16-bit, high-speed oversampled analog-todigital converter (ADC) that offers exceptional dynamic range over a wide bandwidth. The AD9260 is manufactured on an advanced CMOS process. High dynamic range is achieved with an oversampling ratio of 8x through the use of a proprietary technique that combines the advantages of sigma-delta and pipeline converter technologies. The AD9260 is a switched-capacitor ADC with a nominal fullscale input range of 4 V. It offers a differential input with 60 dB of common-mode rejection of common-mode signals. The signal range of each differential input is 1 V centered on a 2.0 V common-mode level. The on-chip decimation filter is configured for maximum performance and flexibility. A series of three half-band FIR filter stages provide 8x decimation filtering with 85 dB of stopband attenuation and 0.004 dB of passband ripple. An onboard digital multiplexer allows the user to access data from the various stages of the decimation filter. The on-chip programmable reference and reference buffer amplifier are configured for maximum accuracy and flexibility. An external reference can also be chosen to suit the user's specific dc accuracy and drift requirements. The AD9260 operates on a single +5 V supply, typically consuming 585 mW of power. A power scaling circuit is provided allowing the AD9260 to operate at power consumption levels as low as 150 mW at reduced clock and data rates. The AD9260 is available in a 44-lead MQFP package and is specified to operate over the industrial temperature range. PRODUCT HIGHLIGHTS The AD9260 is fabricated on a very cost effective CMOS process. High-speed, precision mixed-signal analog circuits are combined with high-density digital filter circuits. The AD9260 offers a complete single-chip 16-bit sampling ADC with a 2.5 MHz output data rate in a 44-lead MQFP. Selectable Internal Decimation Filtering--The AD9260 provides a high-performance decimation filter with 0.004 dB passband ripple and 85 dB of stopband attenuation. The filter is configurable with options for 1x, 2x, 4x, and 8x decimation. Power Scaling--The AD9260 consumes a low 585 mW of power at 16-bit resolution and 2.5 MHz output data rate. Its power can be scaled down to as low as 150 mW at reduced clock rates. Single Supply-- Both of the analog and digital portions of the AD9260 can operate off of a single +5 V supply simplifying system power supply design. The digital logic will also accommodate a single +3 V supply for reduced power. REV. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 (c) Analog Devices, Inc., 2000 AD9260-SPECIFICATIONS CLOCK INPUT FREQUENCY RANGE Parameter--Decimation Factor (N) CLOCK INPUT (Modulator Sample Rate, fCLOCK) OUTPUT WORD RATE (FS = fCLOCK/N) Specifications subject to change without notice AD9260 (8) 1 20 0.125 2.5 AD9260 (4) 1 20 0.250 5 AD9260 (2) 1 20 0.500 10 AD9260 (1) 1 20 1 20 Units kHz min MHz max kHz min MHz max DC SPECIFICATIONS unless otherwise noted, R Parameter--Decimation Factor (N) RESOLUTION INPUT REFERRED NOISE (TYP) 1.0 V Reference 2.5 V Reference1 ACCURACY Integral Nonlinearity (INL) Differential Nonlinearity (DNL) No Missing Codes Offset Error Gain Error2 Gain Error3 TEMPERATURE DRIFT Offset Error Gain Error2 Gain Error3 POWER SUPPLY REJECTION AVDD, DVDD, DRVDD (+5 V 0.25 V) ANALOG INPUT Input Span VREF = 1.0 V VREF = 2.5 V Input (VINA or VINB) Range Input Capacitance INTERNAL VOLTAGE REFERENCE Output Voltage (1 V Mode) Output Voltage Error (1 V Mode) Output Voltage (2.5 V Mode) Output Voltage Error (2.5 V Mode) Load Regulation4 1 V REF 2.5 V REF REFERENCE INPUT RESISTANCE 16 (AVDD = +5 V, DVDD = +3 V, DRVDD = +3 V, fCLOCK = 20 MSPS, VREF = +2.5 V, Input CML = 2.0 V TMIN to TMAX BIAS = 2 k ) AD9260 (8) AD9260 (4) 16 2.4 1.2 (86) 0.75 0.50 16 (0.5) (0.66) (0.7) 2.5 22 7.0 0.06 AD9260 (2) 16 6.0 3.7 (76) 0.75 0.50 16 (0.5) (0.66) (0.7) 2.5 22 7.0 0.06 AD9260 (1) 12 1.3 1.0 (63.2) 0.3 0.25 12 (0.5) (0.66) (0.7) 2.5 22 7.0 0.06 Units Bits min LSB rms typ LSB rms typ (dB typ) LSB typ LSB typ Bits Guaranteed % FSR max (typ @ +25C) % FSR max (typ @ +25C) % FSR max (typ @ +25C) ppm/C typ ppm/C typ ppm/C typ % FSR max 1.40 0.68 (90.6) 0.75 0.50 16 0.9 (0.5) 2.75 (0.66) 1.35 (0.7) 2.5 22 7.0 0.06 1.6 4.0 +0.5 +AVDD - 0.5 10.2 1 14 2.5 35 0.5 2.0 8 1.6 4.0 +0.5 +AVDD - 0.5 10.2 1 14 2.5 35 0.5 2.0 8 1.6 4.0 +0.5 +AVDD - 0.5 10.2 1 14 2.5 35 0.5 2.0 8 1.6 4.0 +0.5 +AVDD - 0.5 10.2 1 14 2.5 35 0.5 2.0 8 V p-p Diff. max V p-p Diff. max V min V max pF typ V typ mV max V typ mV max mV max mV max k -2- REV. B AD9260 Parameter--Decimation Factor (N) POWER SUPPLIES Supply Voltages AVDD DVDD and DRVDD Supply Current IAVDD IDVDD IDRVDD POWER CONSUMPTION AD9260 (8) AD9260 (4) AD9260 (2) AD9260 (1) Units +5 +5.5 +2.7 115 12.5 0.450 613 +5 +5.5 +2.7 115 10.3 0.850 608 +5 +5.5 +2.7 115 6.5 1.7 600 +5 +5.5 +2.7 115 134 2.4 3.5 2.6 585 630 V ( 5%) V max V min mA typ mA max mA typ mA max mA typ mW typ mW max NOTES 1 VINA and VINB Connect to DUT CML. 2 Including Internal 2.5 V reference. 3 Excluding Internal 2.5 V reference. 4 Load regulation with 1 mA load Current (in addition to that required by AD9260). Specifications subject to change without notice. AC SPECIFICATIONS unless otherwise noted, R Parameter--Decimation Factor (N) DYNAMIC PERFORMANCE INPUT TEST FREQUENCY: 100 kHz (typ) Signal-to-Noise Ratio (SNR) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS SNR and Distortion (SINAD) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS Total Harmonic Distortion (THD) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS Spurious Free Dynamic Range (SFDR) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS INPUT TEST FREQUENCY: 500 kHz Signal-to-Noise Ratio (SNR) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS SNR and Distortion (SINAD) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS Total Harmonic Distortion (THD) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS Spurious Free Dynamic Range (SFDR) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS (AVDD = +5 V, DVDD = +3 V, DRVDD = +3 V, fCLOCK = 20 MSPS, VREF = +2.5 V, Input CML = 2.0 V TMIN to TMAX BIAS = 2 k ) AD9260(8) AD9260(4) AD9260(2) AD9260(1) Units 88.5 82.5 87.5 82 -96 -93 100 94 82 78 82 77.5 -96 -98 98 100 74 68 74 69 -97 -96 98 94 63 58 63 58 -98 -98 88 84 dB typ dB typ dB typ dB typ dB typ dB typ dB typ dB typ 86.5 80.5 82.5 86.0 80.0 82.0 -97.0 -90.0 -95.5 99.0 90.0 98 82 77 81 77 -92 -96 92 100 74 68 74 68 -89 -89 91 91 63 58 63 58 -86 -86 88 82 dB typ dB min dB typ dB typ dB min dB typ dB typ dB max dB typ dB typ dB max dB typ REV. B -3- AD9260-SPECIFICATIONS AC SPECIFICATIONS (Continued) Parameter--Decimation Factor (N) DYNAMIC PERFORMANCE (Continued) INPUT TEST FREQUENCY: 1.0 MHz (typ) Signal-to-Noise Ratio (SNR) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS SNR and Distortion (SINAD) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS Total Harmonic Distortion (THD) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS Spurious Free Dynamic Range (SFDR) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS INPUT TEST FREQUENCY: 2.0 MHz (typ) Signal-to-Noise Ratio (SNR) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS SNR and Distortion (SINAD) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS Total Harmonic Distortion (THD) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS Spurious Free Dynamic Range (SFDR) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS INPUT TEST FREQUENCY: 5.0 MHz (typ) Signal-to-Noise Ratio (SNR) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS SNR and Distortion (SINAD) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS Total Harmonic Distortion (THD) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS Spurious Free Dynamic Range (SFDR) Input Amplitude = -0.5 dBFS Input Amplitude = -6.0 dBFS INTERMODULATION DISTORTION fIN1 = 475 kHz, fIN2 = 525 kHz fIN1 = 950 kHz, fIN2 = 1.050 MHz DYNAMIC CHARACTERISTICS Full Power Bandwidth Small Signal Bandwidth (A IN = -20 dBFS) Aperture Jitter Specifications subject to change without notice. AD9260 (8) AD9260 (4) AD9260 (2) AD9260 (1) Units 85 80 84.5 80 -102 -96 105 98 82 76 81 76 -96 -94 98 96 74 68 74 69 -82 -84 83 87 63 58 63 58 -79 -77 80 80 dB typ dB typ dB typ dB typ dB typ dB typ dB typ dB typ 82 76 81 76 -101 -95 104 100 74 68 73 69 -80 -80 80 83 63 58 62 58 -75 -76 78 79 dB typ dB typ dB typ dB typ dB typ dB typ dB typ dB typ 59 57 58 57 -58 -67 59 70 -93 -95 75 75 2 -91 -86 75 75 2 -91 -85 75 75 2 -83 -83 75 75 2 dB typ dB typ dB typ dB typ dB typ dB typ dB typ dB typ dBFS typ dBFS typ MHz typ MHz typ ps rms typ -4- REV. B AD9260 DIGITAL FILTER CHARACTERISTICS Parameter 8x DECIMATION (N = 8) Passband Ripple Stopband Attenuation Passband Stopband Passband/Transition Band Frequency (-0.1 dB Point) (-3.0 dB Point) Absolute Group Delay1 Group Delay Variation Settling Time (to 0.0007%)1 4x DECIMATION (N = 4) Passband Ripple Stopband Attenuation Passband Stopband Passband/Transition Band Frequency (-0.1 dB Point) (-3.0 dB Point) Absolute Group Delay1 Group Delay Variation Settling Time (to 0.0007%)1 2x DECIMATION (N = 2) Passband Ripple Stopband Attenuation Passband Stopband Passband/Transition Band Frequency (-0.1 dB Point) (-3.0 dB Point) Absolute Group Delay1 Group Delay Variation Settling Time (to 0.0007%)1 1x DECIMATION (N = 1) Propagation Delay: tPROP Absolute Group Delay AD9260 0.00125 82.5 0 0.605 x (fCLOCK/20 MHz) 1.870 x (fCLOCK/20 MHz) 18.130 x (fCLOCK/20 MHz) 0.807 x (fCLOCK/20 MHz) 1.136 x (fCLOCK/20 MHz) 13.55 x (20 MHz/fCLOCK) 0 24.2 x (20 MHz/fCLOCK) 0.001 82.5 0 1.24 x (fCLOCK/20 MHz) 3.75 x (fCLOCK/20 MHz) 16.25 x (fCLOCK/20 MHz) 1.61 x (fCLOCK/20 MHz) 2.272 x (fCLOCK/20 MHz) 2.90 x (20 MHz/fCLOCK) 0 5.05 x (20 MHz/fCLOCK) 0.0005 85.5 0 2.491 x (fCLOCK/20 MHz) 7.519 x (fCLOCK/20 MHz) 12.481 x (fCLOCK/20 MHz) 3.231 x (fCLOCK/20 MHz) 4.535 x (fCLOCK/20 MHz) 0.80 x (20 MHz/fCLOCK) 0 1.40 x (20 MHz/fCLOCK) 13 (225 x (20 MHz/fCLOCK)) + tPROP Units dB max dB min MHz min MHz max MHz min MHz max MHz max MHz max s max s max s max dB max dB min MHz min MHz max MHz min MHz max MHz max MHz max s max s max s max dB max dB min MHz min MHz max MHz min MHz max MHz max MHz max s max s max s max ns max ns max NOTES 1 To determine "overall" Absolute Group Delay and/or Settling Time inclusive of delay from the sigma-delta modulator, add Absolute Group Delay and/or Settling Time pertaining to specific decimation mode to the Absolute Group Delay specified in 1 x decimation. Specifications subject to change without notice. REV. B -5- AD9260-Digital Filter Characteristics 0 1.0 -20 NORMALIZED OUTPUT RESPONSE 0.8 0.6 0.4 0.2 MAGNITUDE - dB -40 -60 -80 0 -100 -0.2 -120 0 0.2 0.4 0.6 0.8 FREQUENCY (NORMALIZED TO ) 1.0 -0.4 1.2 0 100 200 300 400 CLOCK PERIODS - RELATIVE TO CLK 500 Figure 1a. 8x FIR Filter Frequency Response Figure 1b. 8x FIR Filter Impulse Response 0 1.0 -20 NORMALIZED OUTPUT RESPONSE 0.8 MAGNITUDE - dB -40 0.6 -60 0.4 -80 0.2 -100 0 -120 0 -0.2 0.2 0.4 0.6 0.8 FREQUENCY (NORMALIZED TO 1.0 ) 1.2 0 10 20 30 40 50 60 70 80 90 100 110 CLOCK PERIODS - RELATIVE TO CLK Figure 2a. 4x FIR Filter Frequency Response Figure 2b. 4x FIR Filter Impulse Response 0 1.0 NORMALIZED OUTPUT RESPONSE -20 0.8 MAGNITUDE - dB -40 0.6 -60 0.4 -80 0.2 -100 0 -120 0 -0.2 0.2 0.4 0.6 0.8 FREQUENCY (NORMALIZED TO ) 1.0 1.2 0 5 10 15 20 CLOCK PERIODS - RELATIVE TO CLK Figure 3a. 2x FIR Filter Frequency Response Figure 3b. 2x FIR Filter Impulse Response -6- REV. B AD9260 Table I. Integer Filter Coefficients for First Stage Decimation Filter (23-Tap Halfband FIR Filter) Table III. Integer Filter Coefficients for Third Stage Decimation Filter (107-Tap Halfband FIR Filter) Lower Coefficient H(1) H(2) H(3) H(4) H(5) H(6) H(7) H(8) H(9) H(10) H(11) H(12) Upper Coefficient H(23) H(22) H(21) H(20) H(19) H(18) H(17) H(16) H(15) H(14) H(13) Integer Value -1 0 13 0 -66 0 224 0 -642 0 2496 4048 Lower Coefficient H(1) H(2) H(3) H(4) H(5) H(6) H(7) H(8) H(9) H(10) H(11) H(12) H(13) H(14) H(15) H(16) H(17) H(18) H(19) H(20) H(21) H(22) H(23) H(24) H(25) H(26) H(27) H(28) H(29) H(30) H(31) H(32) H(33) H(34) H(35) H(36) H(37) H(38) H(39) H(40) H(41) H(42) H(43) H(44) H(45) H(46) H(47) H(48) H(49) H(50) H(51) H(52) H(53) H(54) Upper Coefficient H(107) H(106) H(105) H(104) H(103) H(102) H(101) H(100) H(99) H(98) H(97) H(96) H(95) H(94) H(93) H(92) H(91) H(90) H(89) H(88) H(87) H(86) H(85) H(84) H(83) H(82) H(81) H(80) H(79) H(78) H(77) H(76) H(75) H(74) H(73) H(72) H(71) H(70) H(69) H(68) H(67) H(66) H(65) H(64) H(63) H(62) H(61) H(60) H(59) H(58) H(57) H(56) H(55) Integer Value -1 0 2 0 -2 0 3 0 -3 0 1 0 3 0 -12 0 27 0 -50 0 85 0 -135 0 204 0 -297 0 420 0 -579 0 784 0 -1044 0 1376 0 -1797 0 2344 0 -3072 0 4089 0 -5624 0 8280 0 -14268 0 43520 68508 Table II. Integer Filter Coefficients for Second Stage Decimation Filter (43-Tap Halfband FIR Filter) Lower Coefficient H(1) H(2) H(3) H(4) H(5) H(6) H(7) H(8) H(9) H(10) H(11) H(12) H(13) H(14) H(15) H(16) H(17) H(18) H(19) H(20) H(21) H(22) Upper Coefficient H(43) H(42) H(41) H(40) H(39) H(38) H(37) H(36) H(35) H(34) H(33) H(32) H(31) H(30) H(29) H(28) H(27) H(26) H(25) H(24) H(23) Integer Value 3 0 -12 0 35 0 -83 0 172 0 -324 0 572 0 -976 0 1680 0 -3204 0 10274 16274 NOTE: The composite filter undecimated coefficients (i.e., impulse response) in the 4x decimation mode can be determined by convolving the first stage filter taps with a "zero stuffed" version of the second stage filter taps (i.e., insert one zero between samples). Similarly, the composite filter coefficients in the 8x decimation mode can be determined by convolving the taps of the composite 4x decimation mode (as previously determined) with a "zero stuffed" version of the third stage filter taps (i.e., insert three zeros between samples). REV. B -7- AD9260-SPECIFICATIONS DIGITAL SPECIFICATIONS (AVDD = +5 V, DVDD = +5 V, T Parameter CLOCK AND LOGIC INPUTS High-Level Input Voltage (DVDD = +5 V) (DVDD = +3 V) Low-Level Input Voltage (DVDD = +5 V) (DVDD = +3 V) High-Level Input Current (VIN = DVDD) Low-Level Input Current (VIN = 0 V) Input Capacitance LOGIC OUTPUTS (with DRVDD = 5 V) High-Level Output Voltage (IOH = 50 A) High-Level Output Voltage (IOH = 0.5 mA) Low-Level Output Voltage2 (IOL = 0.3 mA) Low-Level Output Voltage (IOL = 50 A) Output Capacitance LOGIC OUTPUTS (with DRVDD = 3 V) High-Level Output Voltage (IOH = 50 A) Low-Level Output Voltage (IOL = 50 A) 1 MIN to TMAX unless otherwise noted) AD9260 Units +3.5 +2.1 +1.0 +0.9 10 10 5 +4.5 +2.4 +0.4 +0.1 5 +2.4 +0.7 V min V max V min V max A max A max pF typ V min V min V max V max pF typ V min V max NOTES 1 Since CLK is referenced to AVDD, +5 V logic input levels only apply. 2 The AD9260 is not guaranteed to meet V OL = 0.4 V max for standard TTL load of I OL = 1.6 mA. Specifications subject to change without notice. S1 ANALOG INPUT S2 tC tCL tCH INPUT CLOCK tDI DATA OUTPUT tDS tH DAV tOE tDAV tOD READ CS Figure 4a. Timing Diagram tRES-DAV tCLK-DAV INPUT CLOCK RESET DAV Figure 4b. RESET Timing Diagram -8- REV. B AD9260 SWITCHING SPECIFICATIONS Parameters Clock Period Data Available (DAV) Period Data Invalid Data Setup Time Clock Pulsewidth High Clock Pulsewidth Low Data Hold Time RESET to DAV Delay CLOCK to DAV Delay Three-State Output Disable Time Three-State Output Enable Time Specifications subject to change without notice. (AVDD = +5 V, DVDD = +5 V, CL = 20 pF, TMIN to TMAX unless otherwise noted) Symbol tC tDAV tDI tDS tCH tCL tH tRES-DAV tCLK-DAV tOD tOE AD9260 50 tC x Mode 40% tDAV tDAV -tH-tDI 22.5 22.5 3.5 10 15 8 45 Units ns min ns min ns max ns min ns min ns min ns min ns typ ns typ ns typ ns typ ABSOLUTE MAXIMUM RATINGS* ORDERING GUIDE Parameter AVDD DVDD AVSS AVDD DRVDD DRVSS REFCOM CLK, MODE, READ, CS, RESET Digital Outputs VINA, VINB, CML, BIAS VREF SENSE CAPB, CAPT Junction Temperature Storage Temperature Lead Temperature (10 sec) With Respect to Min AVSS DVSS DVSS DVDD DRVSS AVSS AVSS DVSS DRVSS AVSS AVSS AVSS AVSS -0.3 -0.3 -0.3 -6.5 -0.3 -0.3 -0.3 -0.3 -0.3 -0.3 -0.3 -0.3 -0.3 -65 Model Max +6.5 +6.5 +0.3 +6.5 +6.5 +0.3 +0.3 Units V V V V V V V AD9260AS AD9260EB Temperature Range -40C to +85C Package Description 44-Lead MQFP Evaluation Board Package Option* S-44 *S = Metric Quad Flatpack. THERMAL CHARACTERISTICS Thermal Resistance 44-Lead MQFP JA = 53.2C/W JC = 19C/W DVDD + 0.3 V DRVDD + 0.3 V AVDD + 0.3 AVDD + 0.3 AVDD + 0.3 AVDD + 0.3 +150 +150 +300 V V V V C C C *Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum ratings for extended periods may effect device reliability. CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD9260 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. WARNING! ESD SENSITIVE DEVICE REV. B -9- AD9260 DEFINITIONS OF SPECIFICATION INTEGRAL NONLINEARITY (INL) APERTURE JITTER INL refers to the deviation of each individual code from a line drawn from "negative full scale" through "positive full scale." The point used as "negative full scale" occurs 1/2 LSB before the first code transition. "Positive full scale" is defined as a level 1 1/2 LSB beyond the last code transition. The deviation is measured from the middle of each particular code to the true straight line. DIFFERENTIAL NONLINEARITY (DNL, NO MISSING CODES) Aperture jitter is the variation in aperture delay for successive samples and is manifested as noise on the input to the A/D. SIGNAL-TO-NOISE AND DISTORTION (S/N+D, SINAD) RATIO S/N+D is the ratio of the rms value of the measured input signal to the rms sum of all other spectral components below the Nyquist frequency, including harmonics but excluding dc. The value for S/N+D is expressed in decibels. EFFECTIVE NUMBER OF BITS (ENOB) An ideal ADC exhibits code transitions that are exactly 1 LSB apart. DNL is the deviation from this ideal value. Guaranteed no missing codes to 14-bit resolution indicates that all 16384 codes, respectively, must be present over all operating ranges. NOTE: Conventional INL and DNL measurements don't really apply to converters: the DNL looks continually better if longer data records are taken. For the AD9260, INL and DNL numbers are given as representative. ZERO ERROR For a sine wave, SINAD can be expressed in terms of the number of bits. Using the following formula, N = (SINAD - 1.76)/6.02 it is possible to get a measure of performance expressed as N, the effective number of bits. Thus, effective number of bits for a device for sine wave inputs at a given input frequency can be calculated directly from its measured SINAD. TOTAL HARMONIC DISTORTION (THD) The major carry transition should occur for an analog value 1/2 LSB below VINA = VINB. Zero error is defined as the deviation of the actual transition from that point. GAIN ERROR THD is the ratio of the rms sum of the first six harmonic components to the rms value of the measured input signal and is expressed as a percentage or in decibels. SIGNAL-TO-NOISE RATIO (SNR) The first code transition should occur at an analog value 1/2 LSB above negative full scale. The last transition should occur at an analog value 1 1/2 LSB below the nominal full scale. Gain error is the deviation of the actual difference between first and last code transitions and the ideal difference between first and last code transitions. TEMPERATURE DRIFT SNR is the ratio of the rms value of the measured input signal to the rms sum of all other spectral components below the Nyquist frequency, excluding the first six harmonics and dc. The value for SNR is expressed in decibels. SPURIOUS FREE DYNAMIC RANGE (SFDR) The temperature drift for zero error and gain error specifies the maximum change from the initial (+25C) value to the value at TMIN or TMAX. POWER SUPPLY REJECTION SFDR is the difference in dB between the rms amplitude of the input signal and the peak spurious signal. TWO-TONE SFDR The specification shows the maximum change in full scale from the value with the supply at the minimum limit to the value with the supply at its maximum limit. The ratio of the rms value of either input tone to the rms value of the peak spurious component. The peak spurious component may or may not be an IMD product. May be reported in dBc (i.e., degrades as signal level is lowered), or in dBFS (always related back to converter full scale). -10- REV. B AD9260 PIN CONFIGURATION VINB MODE AVDD CAPB AVSS CAPT VINA BIAS CML 44 43 42 41 40 39 38 37 36 35 34 DVSS 1 AVSS 2 DVDD 3 AVDD 4 DRVSS 5 DRVDD 6 CLK 7 READ 8 (LSB) BIT16 9 BIT15 10 BIT14 11 12 13 14 15 16 17 18 19 20 21 22 BIT12 BIT11 BIT10 BIT9 BIT8 BIT7 BIT13 BIT6 BIT5 BIT4 BIT3 PIN 1 IDENTIFIER NC NC 33 REFCOM 32 VREF 31 SENSE 30 RESET AD9260 TOP VIEW (Not to Scale) 29 AVSS 28 AVDD 27 CS 26 DAV 25 OTR 24 BIT1 (MSB) 23 BIT2 NC = NO CONNECT PIN FUNCTION DESCRIPTIONS Pin No. 1 2, 29, 38 3 4, 28, 44 5 6 7 8 9 10-23 24 25 26 27 30 31 32 33 34 35 36 37 39 40, 43 41 42 Name DVSS AVSS DVDD AVDD DRVSS DRVDD CLK READ BIT16 BIT15-BIT2 BIT1 OTR DAV CS RESET SENSE VREF REFCOM MODE BIAS CAPB CAPT CML NC VINA VINB Description Digital Ground. Analog Ground. +3 V to +5 V Digital Supply. +5 V Analog Supply. Digital Output Driver Ground. +3 V to +5 V Digital Output Driver Supply. Clock Input. Part of DSP Interface--Pull Low to Disable Output Bits. Least Significant Data Bit (LSB). Data Output Bit. Most Significant Data Bit (MSB). Out of Range--Set When Converter or Filter Overflows. Data Available. Chip Select (CS): Active LOW. RESET: Active LOW. Reference Amplifier SENSE: Selects REF Level. Input Span Select Reference I/O. Reference Common. Mode Select--Selects Decimation Mode. Power Bias. Noise Reduction Pin--Decouples Reference Level. Noise Reduction Pin--Decouples Reference Level. Common-Mode Level (AVDD/2.5). No Connect (Ground for Shielding Purposes). Analog Input Pin (+). Analog Input Pin (-). REV. B -11- AD9260-Typical Performance Characteristics (AVDD = DVDD = DRVDD = +5.0 V, 4 V Input Span, Differential DC Coupled Input with CML = 2.0 V, fCLOCK = 20 MSPS, Full Bias) 0 100kHz INPUT -20 dB BELOW FULL SCALE 0 100kHz INPUT -20 dB BELOW FULL SCALE 20MHz CLOCK 1 DECIMATION THD: -98dB 20MHz CLOCK 8 DECIMATION THD: -96dB -40 -40 -60 -60 -80 -80 -100 -100 -120 0 0.2 0.4 0.6 0.8 FREQUENCY - MHz 1.0 1.2 -120 0 1 2 3 4 5 6 FREQUENCY - MHz 7 8 9 10 Figure 5. Spectral Plot of the AD9260 at 100 kHz Input, 20 MHz Clock, 8x OSR (2.5 MHz Output Data Rate) Figure 8. Spectral Plot of the AD9260 at 100 kHz Input, 20 MHz Clock, Undecimated (20 MHz Output Data Rate) 0 100kHz INPUT -20 110 -12dBFS/TONE -40 THD: -98dB WORST CASE SPUR - dBFS dB BELOW FULL SCALE 20MHz CLOCK 4 DECIMATION 106 102 -6.5dBFS/TONE 98 -26dBFS/TONE -46dBFS/TONE -60 -80 -100 94 -120 0 0.5 1 1.5 FREQUENCY - MHz 2 2.5 90 0 0.2 0.4 0.6 FREQUENCY - MHz 0.8 1 Figure 6. Spectral Plot of the AD9260 at 100 kHz Input, 20 MHz Clock, 4x OSR (5 MHz Output Data Rate) Figure 9. Dual Tone SFDR vs. Input Frequency (F1 = F2, (F1 - F2, Span = 10% Center Frequency, Mode = 8x) 0 100kHz INPUT -20 dB BELOW FULL SCALE dB BELOW FULL SCALE 0 DUAL-TONE TEST 20MHz CLOCK 2 DECIMATION -20 f1 = 1.0MHz f2 = 975kHz 20MHz CLOCK 8 DECIMATION IM3: -94dB -60 -40 THD: -98dB -40 -60 -80 -80 -100 -100 -120 0 0.5 1 1.5 2 2.5 3 3.5 FREQUENCY - MHz 4 4.5 5 -120 0 0.2 0.4 0.6 0.8 FREQUENCY - MHz 1 1.2 Figure 7. Spectral Plot of the AD9260 at 100 kHz Input, 20 MHz Clock, 2x OSR (10 MHz Output Data Rate) Figure 10. Two-Tone Spectral Performance of the AD9260 Given Inputs at 975 kHz and 1.0 MHz, 20 MHz Clock, 8x Decimation -12- REV. B AD9260 Typical AC Characterization Curves vs. Decimation Mode (AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, Differential DC Coupled Input with CML = 2 V, AIN = 0.5 dBFS Full Bias) 90 85 80 SINAD - dBFS 90 8 MODE 4 80 MODE 8 MODE 4 MODE SINAD - dBFS 85 75 70 65 60 55 50 0.1 2 MODE 75 70 65 60 2 MODE 1 MODE 1 MODE 55 50 0.1 1 INPUT FREQUENCY - MHz 10 1 INPUT FREQUENCY - MHz 10 Figure 11. SINAD vs. Input Frequency (fCLOCK = 20 MSPS)1 Figure 14. SINAD vs. Input Frequency (fCLOCK = 10 MSPS)1 -50 1 -60 MODE -70 -75 -80 -85 1 MODE -70 THD - dBFS THD - dBFS -90 -95 -100 -105 8 MODE 4 MODE 2 MODE -80 2 -90 MODE -100 8 -110 0.1 MODE 4 MODE -110 -115 -120 0.1 1 INPUT FREQUENCY - MHz 10 1 INPUT FREQUENCY - MHz 10 Figure 12. THD vs. Input Frequency (fCLOCK = 20 MSPS) Figure 15. THD vs. Input Frequency (fCLOCK = 10 MSPS) -50 1 -60 MODE -70 -75 -80 -85 1 MODE SFDR - dBFS SFDR - dBFS -70 -90 -95 -100 8 -105 MODE 4 2 MODE MODE -80 2 -90 MODE -110 -100 8 -110 0.1 MODE 10 4 MODE -115 -120 0.1 1 INPUT FREQUENCY - MHz 10 1 INPUT FREQUENCY - MHz Figure 13. SFDR vs. Input Frequency (fCLOCK = 20 MSPS) 1 Figure 16. SFDR vs. Input Frequency (fCLOCK = 10 MSPS) 8x SINAD performance limited by noise contribution of input differential op amp driver. REV. B -13- AD9260 Typical AC Characterization Curves for 8 Mode (AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, Differential DC Coupled Input with CML = 2 V, Full Bias) 90 90 85 -0.5dBFS 80 85 -0.5dBFS 80 -6.0dBFS SINAD - dB SINAD - dB -6.0dBFS 75 75 70 -20dBFS 65 70 -20dBFS 65 60 0.1 INPUT FREQUENCY - MHz 1 60 0.1 INPUT FREQUENCY - MHz 1 Figure 17. SINAD vs. Input Frequency (fCLOCK = 20 MSPS)1 Figure 20. SINAD vs. Input Frequency (fCLOCK = 10 MSPS)1 -70 -75 -20dBFS -80 -85 -70 -75 -80 -20dBFS THD - dB THD - dB -85 -90 -95 -90 -95 -6.0dBFS -0.5dBFS -100 -105 -110 0.1 INPUT FREQUENCY - MHz -100 -105 0.1 INPUT FREQUENCY - MHz -6.0dBFS -0.5dBFS 1 1 Figure 18. THD vs. Input Frequency (fCLOCK = 20 MSPS) Figure 21. THD vs. Input Frequency (fCLOCK = 10 MSPS) 105 105 100 -6.0dBFS SFDR - dBc SFDR - dBc 100 -6.0dBFS 95 95 -0.5dBFS 90 -0.5dBFS 90 85 -20dBFS 85 -20dBFS 80 0.1 INPUT FREQUENCY - MHz 1 80 0.1 INPUT FREQUENCY - MHz 1 Figure 19. SFDR vs. Input Frequency (fCLOCK = 20 MSPS) 1 Figure 22. SFDR vs. Input Frequency (fCLOCK = 10 MSPS) SINAD performance limited by noise contribution of input differential op amp driver. -14- REV. B AD9260 Typical AC Characterization Curves for 4 Mode (AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, Differential DC Coupled Input with CML = 2 V, Full Bias) 90 85 -0.5dBFS 80 90 85 -0.5dBFS 80 -6.0dBFS 75 SINAD - dB 75 70 65 60 55 -6.0dBFS SINAD - dB 70 -20dBFS 65 -20dBFS 50 0.1 1 INPUT FREQUENCY - MHz 10 60 0.1 INPUT FREQUENCY - MHz 1 Figure 23. SINAD vs. Input Frequency (fCLOCK = 20 MSPS) Figure 26. SINAD vs. Input Frequency (fCLOCK = 10 MSPS) -70 -75 -20dBFS -80 -85 -70 -75 -80 -0.5dBFS -85 -0.5dBFS THD - dB THD - dB -20dBFS -90 -95 -6.0dBFS -90 -95 -6.0dBFS -100 -105 -110 0.1 -100 -105 -110 0.1 INPUT FREQUENCY - MHz 1 INPUT FREQUENCY - MHz 10 1 Figure 24. THD vs. Input Frequency (fCLOCK = 20 MSPS) Figure 27. THD vs. Input Frequency (fCLOCK = 10 MSPS) 110 110 105 -0.5dBFS 105 100 -6.0dBFS SFDR - dBc 100 -6.0dBFS 95 SFDR - dBc 95 90 90 -20dBFS 85 85 -20dBFS 80 0.1 -0.5dBFS 80 0.1 1 INPUT FREQUENCY - MHz 10 1 INPUT FREQUENCY - MHz Figure 25. SFDR vs. Input Frequency (fCLOCK = 20 MSPS) Figure 28. SFDR vs. Input Frequency (fCLOCK = 10 MSPS) REV. B -15- AD9260 Typical AC Characterization Curves for 2 Mode (AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, Differential DC Coupled Input with CML = 2 V, Full Bias) 80 80 75 -5.0dBFS 75 -0.5dBFS 70 SINAD - dB 70 SINAD - dB -6.0dBFS -6.0dBFS 65 65 60 60 55 -20dBFS 55 -20dBFS 50 0.1 50 1 INPUT FREQUENCY - MHz 10 0.1 1 INPUT FREQUENCY - MHz 10 Figure 29. SINAD vs. Input Frequency (fCLOCK = 20 MSPS) Figure 32. SINAD vs. Input Frequency (fCLOCK = 10 MSPS) -60 -65 -70 -75 -0.5dBFS -20dBFS -60 -65 -70 -75 THD - dB THD - dB -80 -85 -90 -95 -6.0dBFS -80 -20dBFS -85 -90 -0.5dBFS -95 -6.0dBFS -100 0.1 1.0 INPUT FREQUENCY - MHz 10 -100 0.1 1 INPUT FREQUENCY - MHz 10 Figure 30. THD vs. Input Frequency (fCLOCK = 20 MSPS) Figure 33. THD vs. Input Frequency (fCLOCK = 10 MSPS) 100 100 95 95 -6.0dBFS 90 90 SFDR - dBc SFDR - dBc -6.0dBFS 85 -0.5dBFS 80 -0.5dBFS 85 -20dBFS 80 75 -20dBFS 70 0.1 75 1 INPUT FREQUENCY - MHz 10 70 0.1 1 INPUT FREQUENCY - MHz 10 Figure 31. SFDR vs. Input Frequency (fCLOCK = 20 MSPS) Figure 34. SFDR vs. Input Frequency (fCLOCK = 10 MSPS) -16- REV. B AD9260 Typical AC Characterization Curves for 1 Mode (AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, Differential DC Coupled Input with CML = 2 V, Full Bias) 70 70 65 -0.5dBFS 60 SINAD - dB 65 -0.5dBFS 60 -6.0dBFS 55 SINAD - dB -6.0dBFS 55 50 50 45 -20dBFS 40 0.1 1 INPUT FREQUENCY - MHz 10 45 -20dBFS 40 0.1 1 INPUT FREQUENCY - MHz 10 Figure 35. SINAD vs. Input Frequency (fCLOCK = 20 MSPS) Figure 38. SINAD vs. Input Frequency (fCLOCK = 10 MSPS) -55 -0.5dBFS -60 -20dBFS -65 -70 THD - dB -55 -60 -65 -20dB -6.0dBFS -70 -75 -80 -85 -90 -95 THD - dBc -75 -80 -0.5dBFS -85 -90 -6.0dBFS -95 -100 0.1 1 INPUT FREQUENCY - MHz 10 -100 0.1 1 INPUT FREQUENCY - MHz 10 Figure 36. THD vs. Input Frequency (fCLOCK = 20 MSPS) Figure 39. THD vs. Input Frequency (fCLOCK = 10 MSPS) 100 95 90 85 SDFR - dBc 100 95 -0.5dBFS 90 85 SFDR - dBc -6.0dBFS -0.5dBFS 80 75 70 65 80 75 70 65 60 55 -6.0dBFS -20dBFS -20dBFS 60 55 50 0.1 1 INPUT FREQUENCY - MHz 10 50 0.1 1 INPUT FREQUENCY - MHz 10 Figure 37. SFDR vs. Input Frequency (fCLOCK = 20 MSPS) Figure 40. SFDR vs. Input Frequency (fCLOCK = 10 MSPS) REV. B -17- AD9260 Typical AC Characterization Curves (AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, AIN = -0.5 dBFS, Differential DC Coupled Input with CML = 2 V) 100 95 90 FULL BIAS -60 -65 -70 85 SFDR - dBFS THD - dBc 80 75 70 65 HALF BIAS -75 FIN = 1MHz, 2 -80 -85 -90 FIN = 100kHz, 8 MODE MODE QUARTER BIAS 60 55 50 2 5 10 15 CLOCK FREQUENCY - MHz 20 -95 -100 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 COMMON MODE INPUT LEVEL - Volts 2.8 3.0 Figure 41. SFDR vs. Clock Rate (fIN = 100 kHz in 8x Mode) Figure 44. THD vs. Common-Mode Input Level (CML) -40 100 FULL BIAS -50 80 SFDR - dBFS HALF BIAS FS = 20MHz CMR - dB -60 FS = 10MHz -70 60 QUARTER BIAS 40 20 -80 FS = 5MHz 0 5 10 15 CLOCK FREQUENCY - MHz 20 25 -90 1k 10k 100k 1M INPUT FREQUENCY - Hz 10M 100M Figure 42. SFDR vs. Clock Rate (fIN = 500 kHz in 4x Mode) Figure 45. CMR vs. Input Frequency (VCML = 2 V p-p, 1x Mode) 100 100 FULL BIAS 4V SPAN SNR-8 95 4V SPAN SFDR-2 1.6V SPAN SNR-8 MODE MODE MODE MODE 80 1.6V SPAN SFDR-2 SFDR - dBFS HALF BIAS 60 SFDR - dBFS 90 40 QUARTER BIAS 20 85 80 0 5 10 15 20 CLOCK FREQUENCY - MHz 25 75 0 0.2 0.4 0.6 0.8 1.0 FREQUENCY - MHz 1.2 1.4 1.6 Figure 43. SFDR vs. Clock Rate (fIN = 1.0 MHz in 2x Mode) Figure 46. 4 V vs. 1.6 V Span SNR/SFDR (fCLOCK = 20 MSPS) -18- REV. B AD9260 Additional AC Characterization Curves (AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, AIN = -0.5 dBFS, Differential DC Coupled Input with CML = 2 V, Full Bias, unless otherwise noted) 120 115 110 SFDR - dBFS 105 100 95 90 85 80 -50 120 20MSPS-dBFS FULL BIAS 110 10 MSPS FULL BIAS WORST SPUR - dBc and dBFS 20 MSPS FULL BIAS 100 10MSPS-dBFS HALF BIAS 20MSPS-dBc FULL BIAS 10MSPS-dBc HALF BIAS 90 80 70 60 50 -60 20 MSPS HALF BIAS 10 MSPS HALF BIAS -45 -40 -35 -30 -25 -20 AIN - dBFS -15 -10 -5 0 -50 -40 -30 AIN - dBFS -20 -10 0 Figure 47. Single-Tone SFDR vs. Amplitude (fIN =100 kHz, 8x Mode) Figure 50. Two-Tone SFDR (F1 = 475 kHz, F2 = 525 MHz, 8x Mode) 110 120 FULL BIAS-dBFS 110 WORST SPUR - dBc and dBFS 105 10 MSPS FULL BIAS 100 SFDR - dBFS 100 HALF BIAS-dBFS 90 FULL BIAS-dBc 80 70 60 50 -60 HALF BIAS-dBc 95 20 MSPS FULL BIAS 90 10 MSPS HALF BIAS 85 80 -50 -45 -40 -35 -30 -25 -20 AIN - dBFS -15 -10 -5 0 -50 -40 -30 AIN - dBFS -20 -10 0 Figure 48. Single-Tone SFDR vs. Amplitude (fIN =1.0 MHz, 2x Mode) Figure 51. Two-Tone SFDR (F1 = 0.95 kHz, F2 = 1.05 MHz, 8x Mode 20 MSPS) 110 10 MSPS HALF BIAS 105 10 MSPS FULL BIAS WORST SPUR - dBc and dBFS 120 110 dBFS 100 dBc 90 100 SFDR - dBFS 20 MSPS FULL BIAS 95 80 70 60 50 -60 90 85 80 -50 -45 -40 -35 -30 -25 -20 AIN - dBFS -15 -10 -5 0 -50 -40 -30 AIN - dBFS -20 -10 0 Figure 49. Single-Tone SFDR vs. Amplitude (fIN = 500 kHz, 2x Mode) Figure 52. Two-Tone SFDR (F1 = 1.9 MHz, F2 = 2.1 MHz, 4x Mode 20 MSPS) REV. B -19- AD9260 + - 5B ADC 5B DAC 16 + - 3B ADC 3B DAC 4 + - 3B ADC 3B DAC 4 4B ADC VIN + - INT1 + - INT2 5B DAC1 5B DAC2 MOUT PIPELINE CORRECTION LOGIC 8 LSBs LSB DIFFERENTIATOR SHUFFLE Z-D ++ COUT CONTROL/TEST LOGIC HALF-BAND DECIMATION FILTER STAGE 1 BANDGAP REFERENCE HALF-BAND DECIMATION FILTER STAGE 2 REFERENCE BUFFER HALF-BAND DECIMATION FILTER STAGE 3 OUTPUT BITS Figure 53. Simplified Block Diagram THEORY OF OPERATION The AD9260 utilizes a new analog-to-digital converter architecture to combine sigma-delta techniques with a high-speed, pipelined A/D converter. This topology allows the AD9260 to offer the high dynamic range associated with sigma-delta converters while maintaining very wide input signal bandwidth (1.25 MHz) at a very modest 8x oversampling ratio. Figure 53 provides a block diagram of the AD9260. The differential analog input is fed into a second order, multibit sigma-delta modulator. This modulator features a 5-bit flash quantizer and 5-bit feedback. In addition, a 12-bit pipelined A/D quantizes the input to the 5-bit flash to greater accuracy. A special digital modulation loop combines the output of the 12-bit pipelined A/D with the delayed output of the 5-bit flash to produce the equivalent response of a second order loop with a 12-bit quantizer and 12-bit feedback. The combination of a second order loop and multibit feedback provides inherent stability: the AD9260 is not prone to idle tones or full-scale idiosyncracies sometimes associated with higher order single bit sigmadelta modulators. The output of this 12-bit modulator is fed into the digital decimation filter. The voltage level on the MODE pin establishes the configuration for the digital filter. The user may bring the data out undecimated (at the clock rate), or at a decimation factor of 2x, 4x, or a full 8x. The spectra for these four cases are shown in Figures 5, 6, 7 and 8, all for a 100 kHz full-scale input and 20 MHz clock. The spectra of the undecimated output clearly shows the second order shaping characteristic of the quantization noise as it rises at frequencies above 1.25 MHz. The on-chip decimation filter provides excellent stopband rejection to suppress any stray input signal between 1.25 MHz and 18.75 MHz, substantially easing the requirements on any antialiasing filter for the analog input path. The decimation filters are integrated with symmetric FIR filter structures, providing a linear phase response and excellent passband flatness. The digital output driver register of the AD9260 features both READ and CHIP SELECT pins to allow easy interfacing. The digital supply of the AD9260 is designed to operate over a 2.7 V to 5.25 V supply range, though 3 V supplies are recommended to minimize digital noise on the board. A DATA AVAILABLE pin allows the user to easily synchronize to the converter's decimated output data rate. OUT-OF-RANGE (OTR) indication is given for an overflow in the pipelined A/D converter or digital filters. A RESETB function is provided to synchronize the converter's decimated data and clear any overflow condition in the analog integrators. An on-chip reference and reference buffer are included on the AD9260. The reference can be configured in either a 2.5 V mode (providing a 4 V pk-pk differential input full scale), a 1 V mode (providing a 1.6 V pk-pk differential input full scale), or programmed with an external resistor divider to provide any voltage level between 1 V and 2.5 V. However, optimum noise and distortion performance for the AD9260 can only be achieved with a 2.5 V reference as shown in Figure 46. For users wishing to operate the part at reduced clock frequencies, the bias current of the AD9260 is designed to be scalable. This scaling is accomplished through use of the proper external resistor tied to the BIAS pin: the power can be reduced roughly proportionately to clock frequency by as much as 75% (for clock rates of 5 MHz). Refer to Figures 41-43 and 47-51 for characterization curves showing performance tradeoffs. ANALOG INPUT AND REFERENCE OVERVIEW Figure 54, a simplified model of the AD9260, highlights the relationship between the analog inputs, VINA, VINB and the reference voltage VREF. Like the voltage applied to the top of the resistor ladder in a flash A/D converter, the value VREF defines the maximum input voltage to the A/D converter. An internal reference buffer in the AD9260 scales the reference voltage VREF before it is applied internally to the AD9260 -20- REV. B AD9260 A/D core. The scale factor of this reference buffer is 0.8. Consequently, the maximum input voltage to the A/D core is +0.8 x VREF. The minimum input voltage to the A/D core is automatically defined to be -0.8 x VREF. With this scale factor, the maximum differential input span of 4 V p-p is obtained with a VREF voltage of 2.5 V. A smaller differential input span may be obtained by using a VREF voltage of less than 2.5 V at the expense of ac performance (refer to Figure 46). ANALOG INPUT OPERATION +0.8 VREF VINA + - VINB -0.8 VREF A/D CORE 16 The analog input structure of the AD9260 is optimized to meet the performance requirements for some of the most demanding communication and data acquisition applications. This input structure is composed of a switched-capacitor network that samples the input signal applied to pins VINA and VINB on every rising edge of the CLK pin. The input switched capacitors are charged to the input voltage during each period of CLK. The resulting charge, q, on these capacitors is equal to C x VIN, where C is the input capacitor. The change in charge on these capacitors, delta q, as the capacitors are charged from a previous sample of the input signal to the next sample, is approximated in the following equation, delta q ~ C x deltaVN = C x (VN - VN-2) (4) where VN represents the present sample of the input signal and VN-2 represents the sample taken two clock cycles earlier. The average current flow into the input (provided from an external source) is given in the following equation, I = delta q/T ~ C x (VN - VN-2) x fCLOCK (5) where T represents the period of CLK and fCLOCK represents the frequency of CLK. Equations 4 and 5 provide simplifying approximations of the operation of the analog input structure of the AD9260. A more exact, detailed description and analysis of the input operation is provided below. SS3 SS1 VINA CPA1 CPB1 SS4 ANALOG MODULATOR Figure 54. Simplified Input Model INPUT SPAN The AD9260 is implemented with a differential input structure. This structure allows the common-mode level (average voltage of the two input pins) of the input signal to be varied independently of the input span of the converter over a wide range, as shown in Figure 44. Specifically, the input to the A/D core is the difference of the voltages applied at the VINA and VINB input pins. Therefore, the equation, VCORE = VINA-VINB (1) defines the output of the differential input stage and provides the input to the A/D core. The voltage, VCORE, must satisfy the condition, -0.8 x VREF VCORE +0.8 x VREF where VREF is the voltage at the VREF pin. (2) CS1 SH1 SS2 VINB CPA2 CPB2 CS2 SH2 SH3 INPUT COMPLIANCE RANGE SH4 In addition to the limitations on the differential span of the input signal indicated in Equation 2, an additional limitation is placed on the inputs by the analog input structure of the AD9260. The analog input structure bounds the valid operating range for VINA and VINB. The condition, AVSS +0.5 V < VINA < AVDD - 0.5 V AVSS +0.5 V < VINB < AVDD + 0.5 V (3) Figure 55. Detailed Analog Input Structure where AVSS is nominally 0 V and AVDD is nominally +5 V, defines this requirement. Thus the valid inputs for VINA and VINB are any combination that satisfies both Equations 2 and 3. Note, the clock clamping method used in the differential driver circuit shown in Figure 57 is sufficient for protecting the AD9260 in an undervoltage condition. For additional information showing the relationships between VINA, VINB, VREF and the digital output of the AD9260, see Table V. Refer to Table IV for a summary of the various analog input and reference configurations. Figure 55 illustrates the analog input structure of the AD9260. For the moment, ignore the presence of the parasitic capacitors CPA and CPB. The effects of these parasitic capacitors will be discussed near the end of this section. The switched capacitors, CS1 and CS2, sample the input voltages applied on pins VINA and VINB. These capacitors are connected to input pins VINA and VINB when CLK is low. When CLK rises, a sample of the input signal is taken on capacitors CS1 and CS2. When CLK is high, capacitors CS1 and CS2 are connected to the Analog Modulator. The modulator precharges capacitors CS1 and CS2 to minimize the amount of charge required from any circuit used in combination with the AD9260 to drive input pins VINA and VINB. This reduces the input drive requirements of the analog circuitry driving pins VINA and VINB. The Analog Modulator precharges the voltages across capacitors CS1 and CS2, approximately equal to a delayed version of the input signal. When capacitors CS1 and CS2 are connected to input pins VINA and VINB, the differential charge, Q(n), on these capacitors is given in the following equation, Q(n) = q1 - q2 = CS x VCORE (6) REV. B -21- AD9260 where q1 and q2 are the individual charges stored on capacitors CS1 and CS2 respectively, and CS is the capacitance value of CS1 and CS2. When capacitors CS1 and CS2 are connected to the Analog Modulator during the preceding "precharge" clock phase, the capacitors are precharged equal to an approximation of a previous sample of the input signal. Consequently the differential charge on these capacitors while CLK is high is given in the following equation, Q(n-1) = CS x VCORE(delay) + CS x Vdelta (7) where VCORE(delay) is the value of VCORE sampled during a previous period of CLK, and Vdelta is the sigma-delta error voltage left on the capacitors. Vdelta is a natural artifact of the sigma-delta feedback techniques utilized in the Analog Modulator of the AD9260. It is a small random voltage term that changes every clock period and varies from 0 to 0.05 x VREF. The analog circuitry used to drive the input pins of the AD9260 must respond to the charge glitch that occurs when capacitors CS1 and CS2 are connected to input pins VINA and VINB. This circuitry must provide additional charge, qdelta, to capacitors CS1 and CS2, which is the difference between the precharged value, Q(n-1), and the new value, Q(n), as given in the following equation, Qdelta = Q(n) - Q(n-1) Qdelta = CS x [VCORE-VCORE(delay) + Vdelta] DRIVING THE INPUT Transient Response and CS2 when they are connected to input Pins VINA and VINB. The nonlinear junction capacitance of Cpb1 and Cpb2 cause charge glitch energy that is nonlinearily related to the input signal. Therefore, linear settling is difficult to achieve unless the input source completely settles during one-half period of CLK. A portion of the glitch impulse energy "kicked" back at the source is not linearly related to the input signal. Therefore, the best way to ensure that the input signal settles linearly is to use wide bandwidth circuitry, which settles as completely as possible from the glitch during one-half period of the CLK. The AD9260 utilizes a proprietary clock-boosted boot-strapping technique to reduce the nonlinear parasitic capacitances of the internal CMOS switches. This technique improves the linearity of the input switches and reduces the nonlinear parasitic capacitance. Thus, this technique reduces the nonlinear glitch energy. The capacitance values for the input capacitors and parasitic capacitors for the input structure of the AD9260, as illustrated in Figure 55, are listed as follows. CS = 3.2 pF, Cpa = 6 pF, Cpb = 1 pF (where CS is the capacitance value of capacitors CS1 and CS2, Cpa is the value of capacitors Cpa1 and Cpa2, and Cpb is the value of capacitors Cpb1 and Cpb2). The total capacitance at each input pin is CIN = CS + Cpa + Cpb = 10.2 pF. Input Driver Considerations (8) (9) The charge glitch occurs once at the beginning of every period of the input CLK (falling edge), and the sample is taken on capacitors CS1 and CS2 exactly one-half period later (rising edge). Figure 56 presents a typical input waveform applied to input Pins VINA and VINB of the AD9260. TRACK SAMPLE TRACK SAMPLE TRACK SAMPLE TRACK SAMPLE The optimum noise and distortion performance of the AD9260 can ONLY be achieved when the AD9260 is driven differentially with a 4 V input span . Since not all applications have a signal preconditioned for differential operation, there is often a need to perform a single-ended-to-differential conversion. In the case of the AD9260, a single-ended-to-differential conversion is best realized using a differential op amp driver. Although a transformer will perform a similar function for ac signals, its usefulness is precluded by its inability to directly drive the AD9260 and thus the additional requirement of an active low noise, low distortion buffer stage. Single-Ended-to-Differential Op Amp Driver CLOCK VINA-VINB Figure 56. Typical Input Waveform Figure 56 illustrates the effect of the charge glitch when a source with nonzero output impedance is used to drive the input pins. This source must be capable of settling from the charge glitch in one-half period of the CLK. Unfortunately, the MOS switches used in any CMOS-switched capacitor circuit (including those in the AD9260) include nonlinear parasitic junction capacitances connected to their terminals. Figure 55 also illustrates the parasitic capacitances, Cpa1, Cpb1, Cpa2 and Cpb2, associated with the input switches. Parasitic capacitor Cpa1 and Cpa2 are always connected to Pins VINA and VINB and therefore do not contribute to the glitch energy. Parasitic capacitors Cpb1 and Cpb2, on the other hand, cause a charge glitch that adds to that of input capacitors CS1 There are two single-ended-to-differential op amp driver circuits useful for driving the AD9260. The first circuit, shown in Figure 57, uses the AD8138 and represents the best choice in most applications. The AD8138 is a low-distortion differential ADC driver designed to convert a ground-referenced singleended input signal to a differential output signal with a specified common-mode level for dc-coupling applications. It is capable of maintaining the typical THD and SFDR performance of the AD9260 with only a slight degradation in its noise performance in the 8x mode (i.e., SNR of 85 dB-86 dB). In this application, the AD8138 is configured for unity gain and its common-mode output level is set to 2.5 V (i.e., VREF of the AD9260) to maximize its output headroom while operating from a single supply. Note, single-supply operation has the benefit of not requiring an input protection network for the AD9260 in dc-coupled applications. A simple R-C network at the output is used to filter out high-frequency noise from the AD8138. Recall, the AD9260's small signal bandwidth is 75 MHz, hence any noise falling within the baseband bandwidth of the AD9260 defined by its sample and decimation rate, as well as "images" of its baseband response occurring at multiples of the sample rate, will degrade its overall noise performance. -22- REV. B AD9260 R 499 VIN +5V CS 100pF 499 50 VINA R VCML-VIN 50 50 VINA AD8138 AD9260 VIN R R CF CC 100pF CD 100pF 50 AD9260 50 VINB 499 499 CS 100pF VREF 10 F 0.1 F R R R CF R VCML-VIN 50 CC 100pF VINB AD817 Figure 57. AD8138 Single-Ended Differential ADC Driver CML 0.1 F 1.0 F The second driver circuit, shown in Figure 58, can provide slightly enhanced noise performance relative to the AD8138, assuming low-noise, high-speed op amps are used. This differential op amp driver circuit is configured to convert and level-shift a 2 V p-p single-ended, ground-referenced signal to a 4 V p-p differential signal centered at the common-mode level of the AD9260. The circuit is based on two op amps that are configured as matched unity gain difference amplifiers. The single-ended input signal is applied to opposing inputs of the difference amplifiers, thus providing differential outputs. The common-mode offset voltage is applied to the noninverting resistor leg of each difference amplifier providing the required offset voltage. This offset voltage is derived from the common-mode level (CML) pin of the AD9260 via a low output impedance buffer amplifier capable of driving a 1 F capacitive load. The common-mode offset can be varied over a 1.8 V to 2.5 V span without any serious degradation in distortion performance as shown in Figure 44, thus providing some flexibility in improving output compression distortion from some 5 op amps with limited positive voltage swing. To protect the AD9260 from an undervoltage fault condition from op amps specified for 5 V operation, two 50 series resistors and a diode to AGND are inserted between each op amp output and the AD9260 inputs. The AD9260 will inherently be protected against any overvoltage condition if the op amps share the same positive power supply (i.e., AVDD) as the AD9260. Note, the gain accuracy and common-mode rejection of each difference amplifier in this driver circuit can be enhanced by using a matched thin-film resistor network (i.e., Ohmtek ORNA5000F) for the op amps. Resistor values should be 500 or less to maintain the lowest possible noise. The noise performance of each unity gain differential driver circuit is limited by its inherent noise gain of two. For unity gain op amps ONLY, the noise gain can be reduced from two to one Figure 58. DC-Coupled Differential Driver with Level-Shifting beyond the input signals passband by adding a shunt capacitor, CF, across each op amp's feedback resistor. This will essentially establish a low-pass filter which reduces the noise gain to one beyond the filter's f-3 dB while simultaneously bandlimiting the input signal to f-3 dB. Note, the pole established by this filter can also be used as the real pole of an antialiasing filter. Since the noise contribution of two op amps from the same product family are typically equal but uncorrelated, the total output-referred noise of each op amp will add root-sum square leading to a further 3 dB degradation in the circuit's noise performance. Further out-of-band noise reduction can be realized with the addition of single-ended and differential capacitors, CS and CD. The distortion and noise performance of the two op amps within the signal path are critical in achieving the AD9260's optimum performance. Low noise op amps capable of providing greater than 85 dB THD at 1 MHz while swinging over a 1 V to 3 V range are a rare commodity, yet should only be considered. The AD9632 op amp was found to provide superb distortion performance in this circuit due to its ability to maintain excellent distortion performance over a wide bandwidth while swinging over a 1 V to 3 V range. Since the AD9632 is gain-of-two or greater stable, the use of the noise reduction shunt capacitors discussed above was prohibited thus degrading its noise performance slightly (1 dB-2 dB) when compared to the OPA642. Note, the majority of the AD9260 test and characterization data presented in this data sheet was taken using the AD9632 op amp in this dc coupled driver circuit. This driver circuit is also provided on the AD9260 evaluation board since the AD8138 was unreleased at that time. Table IV. Reference Configuration Summary Reference Operating Mode INTERNAL INTERNAL INTERNAL EXTERNAL Input Span (VINA-VINB) (V p-p) 1.6 4.0 1.6 SPAN 4.0 and SPAN = 1.6 x VREF 1.6 SPAN 4.0 Required VREF (V) 1 2.5 1 VREF 2.5 and VREF = (1+R1/R2) 1 VREF 2.5 Connect SENSE SENSE R1 R2 SENSE VREF To VREF REFCOM VREF and SENSE SENSE and REFCOM AVDD EXT. REF. REV. B -23- AD9260 The outputs of each op amp are ac coupled via a small series resistor and capacitor (i.e., 50 and 0.1 F) to the respective inputs of the AD9260. Similar to the dc coupled driver, further out-of-band noise reduction can be realized with the addition of 100 pF single-ended and differential capacitors, CS and CD. The lower-cutoff frequency of this ac coupled circuit is determined by RC and CC in which RC is tied to the common-mode level pin, CML, of the AD9260 for proper biasing of the inputs. Although the OPA642 was found to provide the lowest overall noise and distortion performance (i.e., 88.8 dB and 96 dB THD @ 100 kHz), the AD8055 (or dual AD8056) suffered only a 0.5 dB to 1.5 dB degradation in overall performance. It is worth noting that given the high-level of performance attainable by the AD9260, special consideration must be given to both the quality of the test equipment and test setup in its evaluation. Common-Mode Level TO A/D 5k CAPT 6.25k 6.25k A2 5k CAPB LOGIC DISABLE A2 -+ 1V A1 7.5k VREF AD9260 SENSE DISABLE A1 LOGIC 5k REFCOM The CML pin is an internal analog bias point used internally by the AD9260. This pin must be decoupled to analog ground with at least a 0.1 F capacitor as shown in Figure 59. The dc level of CML is approximately AVDD/2.5. This voltage should be buffered if it is to be used for any external biasing. Note: the common-mode voltage of the input signal applied to the AD9260 need not be at the exact same level as CML. While this level is recommended for optimal performance, the AD9260 is tolerant of a range of input common-mode voltages around AVDD/2.5. CML 0.1 F Figure 60. Simplified Reference AD9260 Figure 59. CML Decoupling REFERENCE OPERATION The AD9260 contains an onboard bandgap reference and internal reference buffer amplifier. The onboard reference provides a pin-strappable option to generate either a 1 V or 2.5 V output. With the addition of two external resistors, the user can generate reference voltages other than 1 V and 2.5 V. Another alternative is to use an external reference for designs requiring enhanced accuracy and/or drift performance. See Table IV for a summary of the pin-strapping options for the AD9260 reference configurations. Note, the optimum noise and distortion can only be achieved with a 2.5 V reference. Figure 60 shows a simplified model of the internal voltage reference of the AD9260. A pin-strappable reference amplifier buffers a 1 V fixed reference. The output from the reference amplifier, A1, appears on the VREF pin and MUST be decoupled with 0.1 F and 10 F capacitor to REFCOM. The voltage on the VREF pin determines the full-scale input span of the A/D. This input span equals: Full-Scale Input Span = 1.6 x VREF The voltage appearing at the VREF pin, as well as the state of the internal reference amplifier, A1, are determined by the voltage appearing at the SENSE pin. The logic circuitry contains two comparators that monitor the voltage at the SENSE pin. The comparator with the lowest set point (approximately 0.3 V) controls the position of the switch within the feedback path of A1. If the SENSE pin is tied to REFCOM, the switch is connected to the internal resistor network, thus providing a VREF of 2.5 V. If the SENSE pin is tied to the VREF pin via a short or resistor, the switch is connected to the SENSE pin. A short will provide a VREF of 1.0 V while an external resistor network will provide an alternative VREF SPAN between 1.0 V and 2.5 V. The external resistor network may, for example, be implemented as a resistor divider circuit. This divider circuit could consist of a resistor (R1) connected between VREF and SENSE and another resistor (R2) connected between SENSE and REFCOM. The other comparator controls internal circuitry that will disable the reference amplifier if the SENSE pin is tied to AVDD. Disabling the reference amplifier allows the VREF pin to be driven by an external voltage reference. The reference buffer circuit, level shifts the reference to an appropriate common-mode voltage for use by the internal circuitry. The on-chip buffer provides the low impedance necessary for driving the internal switched capacitor circuits and eliminates the need for an external buffer op amp. The actual reference voltages used by the internal circuitry of the AD9260 appear on the CAPT and CAPB pins. If VREF is configured for 2.5 V, thus providing a 4 V full-scale input span, the voltages appear at CAPT and CAPB are 3.0 V and 1.0 V respectively. For proper operation when using the internal or an external reference, it is necessary to add a capacitor network to decouple the CAPT and CAPB pins. Figure 61 shows the recommended decoupling network. This capacitive network performs the following three functions: (1) along with the reference amplifier, A2, it provides a low source impedance over a large frequency range to drive the A/D internal circuitry, (2) it provides the necessary compensation for A2, and (3) it bandlimits the noise contribution from the reference. The turn-on time of the reference voltage appearing between CAPT and CAPB is approximately 15 ms and should be evaluated in any powerdown mode of operation. REV. B -24- AD9260 AD9260 VREF 0.1 F + 10 F SENSE REFCOM CAPB 0.1 F CAPT 0.1 F + 10 F 0.1 F state. Table VI indicates the relationship between the CS and READ pins and the state of Pins Bit 1-Bit 16. Table VI. CS and READ Pin Functionality CS Low Low High High DAV PIN READ Low High Low High Condition of Data Output Pins Data Output Pins in Hi-Z State ADC Data on Output Pins Data Output Pins in Hi-Z State Data Output Pins in Hi-Z State Figure 61. Recommended Reference Decoupling Network DIGITAL INPUTS AND OUTPUTS Digital Outputs The AD9260 output data is presented in a twos complement format. Table V indicates the output data formats for various input ranges and decimation modes. A straight binary output data format can be created by inverting the MSB. Table V. Output Data Format Input (V) Condition (V) Digital Output 1000 0000 0000 0000 1000 0000 0000 0000 0000 0000 0000 0000 0111 1111 1111 1111 0111 1111 1111 1111 1000 0001 0001 1100 1000 0001 0000 1100 0000 0000 0000 0000 0111 1110 1110 0011 0111 1110 1110 0011 1000 0000 0100 0001 1000 0000 0100 0001 0000 0000 0000 0000 0111 1111 1011 1110 0111 1111 1011 1110 8 Decimation Mode VINA-VINB < -0.8 x VREF VINA-VINB = -0.8 x VREF VINA-VINB = 0 VINA-VINB = +0.8 x VREF - 1 LSB VINA-VINB >= + 0.8 x VREF 4 Decimation Mode VINA-VINB < -0.825 x VREF VINA-VINB = -0.825 x VREF VINA-VINB = 0 VINA-VINB = +0.825 x VREF - 1 LSB VINA-VINB >= + 0.825 x VREF 2 Decimation Mode VINA-VINB < -0.825 x VREF VINA-VINB = -0.825 x VREF VINA-VINB = 0 VINA-VINB = +0.825 x VREF - 1 LSB VINA-VINB >= + 0.825 x VREF The DAV pin indicates when the output data of the AD9260 is valid. Digital output data is updated on the rising edge of DAV. The data hold time (tH) is dependent on the external loading of DAV and the digital data output pins (BIT1-BIT16) as well as the particular decimation mode. The internal DAV driver is sized to be larger than the drivers pertaining to the digital data outputs to ensure that rising edge of DAV occurs before the data transitions under similar loading conditions (i.e., fanout) regardless of mode. Note that minimum data hold (tH) of 3.5 ns is specified in the Figure 4 timing diagram from the 50% point of DAV's rising edge to the 50% of data transition using a capacitive load of 20 pF for DAV and BIT1-BIT16. Applications interfacing to TTL logic and/or having larger capacitive loading for DAV than BIT1-BIT16 should consider latching data on the falling edge of DAV since the falling edge of DAV occurs well after the data has transitioned in the case of the 2x, 4x and 8x modes. The duty cycle of DAV is approximately 50% and it remains active independent of CS and READ. RESET PIN The slight different full-scale input voltage conditions and their corresponding digital output code for the 4x and 2x decimation modes can be attributed to the different digital scaling factors applied to each of the AD9260's FIR decimation stages for filter optimization purposes. Thus, a + full-scale reading of 0111 1111 1111 1111 and - full-scale reading of 1000 0000 0000 0000 is unachievable in the 2x and 4x decimation mode. As a result, a digital overrange condition can never exist in the 2x and 4x decimation mode and thus OTR being set high indicates an overrange condition in the analog modulator. The output data format in 1x decimation differs from that in 2x, 4x and 8x decimation modes. In 1x decimation mode the output data remains in a twos complement format, but the digital numbers are scaled by a factor of 7/128. This factor of 7/128 is the product of an internal scale factor of 7/8 in the analog modulator and a 1/16 scale factor caused by LSB justification of the 12-bit modulator data. CS AND READ PINS The RESET pin is an asynchronous digital input that is active low. Upon asserting RESET low, the clocks in the digital decimation filters are disabled, the DAV pin goes low and the data on the digital output data pins (Bit 1-Bit 16) is invalid. In addition, the analog modulator in the AD9260 and internal clock dividers used in the decimation filters are reset and will remain reset as long as RESET is maintained low. In the 2x, 4x, or 8x mode, the RESET must remain low for at least a clock period to ensure all the clock dividers and analog modulator are reset. Upon bringing RESET high, the internal clock dividers will begin to count again on the next falling edge of CLK and DAV will go high approximately 15 ns after this falling edge, resuming normal operation. Refer to Figure 4b for a timing diagram. The state of the internal decimation filters in the AD9260 remains unchanged when RESET is asserted low. Consequently, when RESET is pulsed low, this resets the analog modulator but does not clear all the data in the digital filters. The data in the filters is corrupted by the effect of resetting the analog modulator (this causes an abrupt change at the input of the digital filter and this change is unrelated to the signal at the input of the A/D converter). Similarly, in multiplexed applications in which the input of the A/D converters sees an abrupt change, the data in the analog modulator and digital filter will be corrupted. For this reason, following a pulse on the RESET pin, or change in channels (i.e., multiplexed applications only), the decimation filters must be flushed of their data. These filters have a memory length, hence delay, equal to the number of filter taps times the clock rate of the converter. This memory length may be -25- The CS and READ pins control the state of the output data pins (BIT1-BIT16) on the AD9260. The CS pin is active low and the READ pin is active high. When CS and READ are both active the ADC data is driven on the output data pins, otherwise the output data pins are in a high-impedance (Hi-Z) REV. B AD9260 interpreted in terms of a number of samples stored in the decimation filter. For example, if the part is in 8x decimation mode, the delay is 321/fCLOCK. This corresponds to 321 samples stored in the decimation filter. These 321 samples must be flushed from the AD9260 after RESET is pulsed high prior to reusing the data from the AD9260. That is, the AD9260 should be allowed to clock for 321 samples as the corrupted data is flushed from the filters. If the part is in 4x or 2x decimation mode, then the relatively smaller group delays of the 4x and 2x decimation filters result fewer samples that must be flushed from the filters (108 samples and 23 samples respectively). In 2x, 4x or 8x mode, RESET may be used to synchronize multiple AD9260s clocked with the same clock. The decimation filters in the AD9260 are clocked with an internal clock divider. The state of this clock divider determines when the output data becomes available (relative to CLK). In order to synchronize multiple AD9260s clocked with the same clock, it is necessary that the clock dividers in each of the individual AD9260s are all reset to the same state. When RESET is asserted low, these clock dividers are cleared. On the next falling edge of CLK following the rising edge of RESET, the clock dividers begin counting and the clock is applied to the digital decimation filters. OTR PIN The second out-of-range detector is placed at the output of the stage three decimation filter and detects whether the low pass filtered data has exceeded full scale. When this occurs, the filter output data is hard limited to full scale. The OTR signal is a logical OR function of the signals from these two internal outof-range detectors. If either of these detectors produces an outof-range signal, the OTR pin goes high and the data may be seriously corrupted. If the AD9260 is used in a system that incorporates automatic gain control (AGC), the OTR signal may be used to indicate that the signal amplitude should be reduced. This may be particularly effective for use in maximizing the signal dynamic range if the signal includes high-frequency components that occasionally exceed full scale by a small amount. If, on the other hand, the signal includes large amplitude low frequency components that cause the digital filters to overrange, this may cause the low pass digital filter to overrange. In this case the data may become seriously corrupted and the digital filters may need to be flushed. See the RESET pin function description above for an explanation of the requirements for flushing the digital filters. OTR should be sampled with the falling edge of CLK. This signal is invalid while CLK is HIGH. MODE OPERATION The OTR pin is a synchronous output that is updated each CLK period. It indicates that an overrange condition has occurred within the AD9260. Ideally, OTR should be latched on the falling edge of CLK to ensure proper setup-and-hold time. However, since an overrange condition typically extends well beyond one clock cycle (i.e., does not toggle at the CLK rate). OTR typically remains high for more than a clock cycle, allowing it to be successfully detected on the rising edge of CLK or monitored asynchronously. An overrange condition must be carefully handled because of the group delays in the low-pass digital decimation filters in the output stages of the AD9260. When the input signal exceeds the full-scale range of the converter, this can have a variety of effects upon the operation of the AD9260, depending on the duration and amplitude of this overrange condition. A short duration overrange condition (<< filter group delay) may cause the analog modulator to briefly overrange without causing the data in the low pass digital filters to exceed full scale. The analog modulator is actually capable of processing signals slightly (3%) beyond the full-scale range of the AD9260 without internally clipping. A long duration overrange condition will cause the digital filter data to exceed full scale. For this reason, the OTR signal is generated using two separate internal out-ofrange detectors. The first of these out-of-range detectors is placed at the output of the analog modulator and indicates whether the modulator output signal has extended 3% beyond the full-scale range of the converter. If the modulator output signal exceeds 3% beyond full scale, the digital data is hard-limited (i.e., clipped) to a number that is 3% larger than full scale. Due to the delay of the switched capacitor analog modulator, the OTR signal is delayed 3 1/2 clock cycles relative to the clock edge in which the overranged analog input signal was sampled. The Mode Select Pin (MODE) allows the user to select one of four available digital filter modes using a single pin. Each mode configures the internal decimation filter to decimate at: 1x, 2x, 4x or 8x. Refer to Table VII for mode pin ranges. The mode selection is performed by using a set of internal comparators, as illustrated in Figure 62, so that each mode corresponds to a voltage range on the input of the MODE pin. The output of the comparators are fed into encoding logic where, on the falling edge of the clock, the encoded data is latched. Table VII. Recommended Mode Pin Ranges and Configurations Mode Pin Range 0 V-0.5 V 0.5 V-1.5 V 1.5 V-3.0 V 3.0 V-5.0 V BIAS PIN OPERATION Typical Mode Pin GND VREF/2 CML AVDD Decimation Mode 8x 2x 4x 1x The Bias Select Pin (BIAS) gives the user, who is able to operate the AD9260 at a slower clock rate, the added flexibility of running the device in a lower, power consumption mode when it is clocked at less than 20 MHz. This is accomplished by scaling the bias current of the AD9260 as illustrated in Figure 63. The bias amplifier drives a source follower and forces 1 V across REXT, which sets the bias current. This effectively adjusts the bias current in the modulator amplifiers and FLASH preamplifiers. When a large value of REXT is used, a smaller bias current is available to the internal amplifier circuitry. As a result these amplifiers need more time to settle, thus dictating the use of a slower clock as the power is reduced. Refer to the characterization curves shown in Figures 41-48 revealing the performance tradeoffs. -26- REV. B AD9260 The scaling is accomplished by properly attaching an external resistor to the BIAS pin of the AD9260 as shown in Table IX. REXT is normally 2 k for a clock speed of 20 MHz and scales inversely with clock rate. Because BIAS is an external pin, minimization of capacitance to this pin is recommended in order to prevent instability of the bias pin amplifier. AVDD 130 FULL BIAS-2k 110 IAVDD - mA 90 HALF BIAS-4k 70 4R 50 3R MODE PIN QUARTER BIAS-8k 30 5 ENCODER ENCODED MODE 10 15 SAMPLE RATE - MSPS 20 LATCH 2R Figure 64. IAVDD vs. Sample Rate (AVDD = +5 V, Mode 1x-4x) 16 4 14 12 IDVDD/IDRVDD - mA 8 CLOCK R AVSS 1 10 2 8 6 4 Figure 62. Simplified Mode Pin Circuitry BIAS CURRENT 1V 2 0 5 BIAS PIN REXT 10 15 SAMPLE RATE - MSPS 20 Figure 65a. IDVDD/IDRVDD vs. Sample Rate (DVDD = DRVDD = 3 V, fIN = 1 MHz) 30 4 25 8 Figure 63. Simplified Bias Pin Circuitry POWER DISSIPATION CONSIDERATIONS The power dissipation of the AD9260 is dependent on its application-specific configuration and operating conditions. The analog power dissipation as shown in Figure 64 is primarily a function of its power bias setting and sample rate. It remains insensitive to the particular input waveform being digitized or digital filter MODE setting. The digital power dissipation is primarily a function of the digital supply setting (i.e., +3 V to +5 V), the sample rate and, to a lesser extent, the MODE setting and input waveform. Figures 65a and 65b show the total current dissipation of the "combined" digital (DVDD) and digital driver supply (DRVDD) for +3 V and +5 V supplies. Note, DVDD and DRVDD are typically derived from the same supply bus since no degradation in performance results. A 1 MHz fullscale sine wave was used to ensure maximum digital activity in the digital filters and the digital drivers had a fanout of one. Note also that a twofold decrease in digital supply current results when the digital supply is reduced form +5 V to +3 V. IDVDD/IDRVDD - mA 20 1 15 2 10 5 0 5 10 15 SAMPLE RATE - MSPS 20 Figure 65b. IDVDD/IDRVDD vs. Sample Rate (DVDD = DRVDD = 5 V, fIN = 1 MHz) REV. B -27- AD9260 Digital Output Driver Considerations (DRVDD) The AD9260 output drivers can be configured to interface with +5 V or 3.3 V logic families by setting DRVDD to +5 V or 3.3 V respectively. The AD9260 output drivers in each mode are appropriately sized to provide sufficient output current to drive a wide variety of logic families. However, large drive currents tend to cause glitches on the supplies and may affect SINAD performance. Applications requiring the AD9260 to drive large capacitive loads or large fanout may require additional decoupling capacitors on DRVDD. The addition of external buffers or latches helps reduce output loading while providing effective isolation from the databus. Clock Input and Considerations These characteristics result in both a reduction of electromagnetic interference (EMI) and an overall improvement in performance. It is important to design a layout that prevents noise from coupling onto the input signal. Digital signals should not be run in parallel with input signal traces and should be routed away from the input circuitry. While the AD9260 features separate analog and digital ground pins, it should be treated as an analog component. The AVSS, DVSS and DRVSS pins must be joined together directly under the AD9260. A solid ground plane under the A/D is acceptable if the power and ground return currents are managed carefully. Alternatively, the ground plane under the A/D may contain serrations to steer currents in predictable directions where cross-coupling between analog and digital would otherwise be unavoidable. The AD9260/EB ground layout, shown in Figure 76, depicts the serrated type of arrangement. The analog and digital grounds are connected by a jumper below the A/D. Analog and Digital Supply Decoupling The AD9260 internal timing uses the two edges of the clock input to generate a variety of internal timing signals. The clock input must meet or exceed the minimum specified pulse width high and low (tCH and tCL) specifications for the given A/D as defined in the Switching Specifications at the beginning of the data sheet to meet the rated performance specifications. For example, the clock input to the AD9260 operating at 20 MSPS may have a duty cycle between 45% to 55% to meet this timing requirement since the minimum specified tCH and tCL is 22.5 ns. For clock rates below 20 MSPS, the duty cycle may deviate from this range to the extent that both tCH and tCL are satisfied. All high-speed high-resolution A/Ds are sensitive to the quality of the clock input. The degradation in SNR at a given full-scale input frequency (fIN) due to only aperture jitter (tA) can be calculated with the following equation: SNR = 20 log10 [1/(2 fIN tA)] In the equation, the rms aperture jitter, tA, represents the rootsum square of all the jitter sources which include the clock input, analog input signal, and A/D aperture jitter specification. For example, if a 500 kHz full-scale sine wave is sampled by an A/D with a total rms jitter of 15 ps, the SNR performance of the A/D will be limited to 86.5 dB. The clock input should be treated as an analog signal in cases where aperture jitter may affect the dynamic range of the AD9260. In fact, the CLK input buffer is internally powered from the AD9260's analog supply, AVDD. Thus the CLK logic high and low input voltage levels are +3.5 V and +1.0 V, respectively. Supplies for clock drivers should be separated from the A/D output driver supplies to avoid modulating the clock signal with digital noise. Low jitter crystal controlled oscillators make the best clock sources. If the clock is generated from another type of source (by gating, dividing, or other method), it should be retimed by the original clock at the last step. GROUNDING AND DECOUPLING Analog and Digital Grounding The AD9260 features separate analog, digital, and driver supply and ground pins, helping to minimize digital corruption of sensitive analog signals. Figure 66 shows the power supply rejection ratio vs. frequency for a 200 mV p-p ripple applied to AVDD, DVDD, and DAVDD. 90 85 DVDD & DRVDD 80 75 PSRR - dBFS 70 65 60 55 AVDD 50 45 40 1 10 100 FREQUENCY - kHz 1000 10000 Figure 66. AD9260 PSRR vs. Frequency (8 x Mode) In general, AVDD, the analog supply, should be decoupled to AVSS, the analog common, as close to the chip as physically possible. Figure 67 shows the recommended decoupling for the analog supplies; 0.1 F ceramic chip capacitors should provide adequately low impedance over a wide frequency range. Note that the AVDD and AVSS pins are co-located on the AD9260 Proper grounding is essential in any high-speed, high-resolution system. Multilayer printed circuit boards (PCBs) are recommended to provide optimal grounding and power schemes. The use of ground and power planes offers distinct advantages: 1. The minimization of the loop area encompassed by a signal and its return path. 2. The minimization of the impedance associated with ground and power paths. 3. The inherent distributed capacitor formed by the power plane, PCB insulation, and ground plane. -28- 4 AVDD 0.1 F 3 AVSS AVDD 44 0.1 F AVSS 38 AD9260 28 AVDD 0.1 F 29 AVSS Figure 67. Analog Supply Decoupling REV. B AD9260 to simplify the layout of the decoupling capacitors and provide the shortest possible PCB trace lengths. The AD9260/EB power plane layout, shown in Figure 77 depicts a typical arrangement using a multilayer PCB. The digital activity on the AD9260 chip falls into two general categories: digital logic, and output drivers. The internal digital logic draws surges of current, mainly during the clock transitions. The output drivers draw large current impulses while the output bits are changing. The size and duration of these currents are a function of the load on the output bits: large capacitive loads are to be avoided. Note that the digital logic of the AD9260 is referenced DVDD while the output drivers are referenced to DRVDD. Also note that the SNR performance of the AD9260 remains independent of the digital or driver supply setting. The decoupling shown in Figure 68, a 0.1 F ceramic chip capacitor, is appropriate for a reasonable capacitive load on the digital outputs (typically 20 pF on each pin). Applications involving greater digital loads should consider increasing the digital decoupling proportionally, and/or using external buffers/ latches. 3 DVDD 0.1 F 1 DVSS DRVDD 6 0.1 F DRVSS 5 SAMPLING CLOCK GENERATOR Speed Design Techniques seminar book, which is available at www.analog.com/support/frames/lin_frameset.hml. INSERT 5/3 VOLT LINEAR REGULATOR FOR 3 OR 3.3V DIGITAL OPERATION FERRITE BEAD CORE* VA 10 F DVDD 0.1 F DVSS DRVSS DRVDD 0.1 F VD AD9260 AVDD 0.1 F AVSS BITS 1-16, DAV AVDD 0.1 F AVSS CLK AVDD 0.1 F AVSS BUFFER LATCH AD9260 Figure 69. Figure 68. Digital Supply Decoupling A complete decoupling scheme will also include large tantalum or electrolytic capacitors on the PCB to reduce low-frequency ripple to negligible levels. Refer to the AD9260/EB schematic and layouts in Figures 73-77 for more information regarding the placement of decoupling capacitors. An alternative layout and decoupling scheme is shown in Figure 69. This layout and decoupling scheme is well suited for applications in which multiple AD9260s are located on the same PC board and/or the AD9260 is part of a multicard mixed signal system in which grounds are tied back at the system supplies (i.e., star ground configuration). In this case, the AD9260 is treated as an analog component in which its analog (i.e., AVDD) and digital (DVDD and DRVDD) supplies are derived from the systems +5 V analog supply and all of the AD9260's ground pins are tied directly to the analog ground plane which resides directly underneath the IC. Referring to Figure 69, each supply pin is directly decoupled to their respective ground pin or analog ground plane via a ceramic 0.1 F chip capacitor. Surface mount ferrite beads are used to isolate the analog (AVDD), digital (DVDD), and driver supplies (DRVDD) of the AD9260 from the +5 V power buss. Properly selected ferrite beads can provide more than 40 dB of isolation from high-frequency switching transients originating from AD9260 supply pins. Further noise immunity from noise is provided by the inherent power-supply rejection of the AD9260 as shown in Figure 64. If digital operation at 3 V is desirable for power savings and or to provide for a 3 V digital logic interface, a 5 V to 3 V linear regulator can be used to drive DVDD and/or DRVDD. A more complete discussion on this layout and decoupling scheme can be found in Chapter 7, pages 7-27 through 7-55 of the High AD9260 EVALUATION BOARD GENERAL DESCRIPTION The AD9260 Evaluation Board is designed to provide an easy and flexible method of exercising the AD9260 and demonstrate its performance to data sheet specifications. The evaluation board is fabricated in four layers: the component layer; the ground layer; the power layer and the solder layer. The board is clearly labeled to provide easy identification of components. Ample space is provided near the analog and clock inputs to provide additional or alternate signal conditioning. FEATURES AND USER CONTROL * Jumper Controlled Mode/OSR Selection: The choice of Mode/OSR can easily be varied by jumping either JP1, JP2, JP3 or JP4 as illustrated in Figure 71 within the Mode/OSR Control Block. To obtain the desired mode refer to Table VIII. Table VIII. AD9260 Evaluation Board Mode Select Mode/OSR 1x 2x 4x 8x Connect Jumper JP4 JP2 JP3 JP1 * Selectable Power Bias: The power consumption of the AD9260 can be scaled down if the user is able to operate the device at a lower clock frequency. As illustrated in Figure 71, pin cups are provided for the external resistor (R2) tied to the BIAS pin of the AD9260. Table IX defines the recommended resistance for a given clock speed to obtain the desired power consumption. REV. B -29- AD9260 U5 1 NC 2.5/3V 7 AD780R +VIN 2 NC 6 3 TEMP VOUT 5 4 GNDS TRIM 8 VCC2 R10 1k 1KPOT C18 0.1 F AGND C19 0.1 F R3 15k R12 15k C14 0.1 F JP10 R11 49.9 AD817R U6 Q1 2N2222 R8 390 AGND VCC2 AGND + C12 0.1 F VREFEXT 1V R9 1k C13 10 F R4 10k R13 10k C17 10 F + C15 0.1 F Figure 70. Evaluation Board External Reference Circuitry Table IX. Evaluation Board Recommended Resistance Value for External Bias Resistor Resistor Value 2 k 4 k 8 k 16 k Clock Speed (max) 20 MHz 10 MHz 5 MHz 2.5 MHz Power Consumption 585 mW 325 mW 200 mW 150 mW The external reference circuitry, is illustrated in Figure 70. By connecting or disconnecting JP10, the external reference can be configured for either 1.0 V or 2.5 V. That is, by connecting JP10, the external reference will be configured to provide a 2.5 V reference. By leaving JP10 open, the external reference will be configured to provide a 1.0 V reference. * Flexible DC or AC Coupled External Clock Inputs: As illustrated in Figure 71, the AD9260 Evaluation Board is designed to allow the user the flexibility of selecting how to connect the external clock source. It is also equipped with a playpen area for experimenting with optional clock drivers or crystals. * Selecting DC or AC Coupled External Clock: DC Coupled: To directly drive the clock externally via the CLKIN connector, connect JP11 and disconnect JP12. Note: 50 terminated by R27. AC Coupled: To ac couple the external clock and level shift it to midsupply, connect JP12 and disconnect JP11. Note: 50 terminated by R27. * Flexible Input Signal Configuration Circuitry: The AD9260 Evaluation Board's Input Signal Configuration Block is illustrated in Figure 72. It is comprised of an input signal summing amplifier (U7), a variable input signal commonmode generator (U10) and a pair of amplifiers (U8 and U9) that configure the input into a differential signal and drive it, through a pair of isolation resistors, into the input pins of AD9260. The user can either input a signal or dual signal into the evaluation board via the two SMA connectors (J6 and J7) labeled IN-1 or IN-2. The user should refer to the Driving the Input section of the data sheet for a detailed explanation of how the inputs are to be driven and what amplifier requirements are recommended. * Selecting Single or Dual Signal Input: The input amplifier (U7) can either be configured as a dual input signal inverting summer or a single tone inverting buffer. This flexibility will allow for slightly better noise performance in the single tone mode due to the inherent noise gain difference in the two amplifier configurations. An optional feedback capacitor (C9) was added to allow the user additional out-of band filtering of the input signal if needed. -30- REV. B * Data Interfacing Controls: The data interfacing controls (RESETB, CSB, READ, DAV) are all accessible via SMA connectors (J2-J5) as illustrated in Figure 71 within the data interfacing control block. The RESETB, CSB and READ connections are each supplied with two sets or resistor pin cups to allow the user to pull-up or pull-down each signal to a fixed state. R5, R6 and R30 will terminate to ground, while R7, R28 and R29 terminate to DRVDD. The DAV and OTR signals are also directly connected to the data output connector P1. All interfacing controls are buffered through the CMOS line driver 74HC541. * Buffered Output Data: The twos complement output data is buffered through two CMOS noninverting bus transceivers (U2 and U3) and made available at pin connector P1 as illustrated in Figure 71 within the data output block. * Jumper Controlled Reference Source: The choice of reference for the AD9260 can easily be varied between 1.0 V, 2.5 V or external, by using Jumpers JP5, JP6, JP7 and JP9 as illustrated in Figure 71 within the reference configuration block. To obtain the desired reference see Table X. Table X. Evaluation Board Reference Pin Configuration Reference Voltage 2.5 V 1.0 V External Connect Jumper JP7 JP6 JP5, JP9 and JP10 Input Voltage (pk-pk FS) 4.0 V 1.6 V 4.0 V AD9260 For two-tone input signals: The user would leave jumpers (JP8) connected and use IN-1 and IN-2 (J7 and J6) as the connectors for the input signals. For signal tone input signal: The user would remove jumper (JP8) and use only IN-1 as the input signal connector. * Selectable Input Signal Common-Mode Level Source: The input signal's common-mode level (CML) can be set by U10. To use the Input CML generated by U10: Disconnect jumper JP13 and Connect resistors RX3 and RX4. The CML generated by U10 is variable and adjustable using the 1 k trimpot R35. SHIPMENT CONFIGURATION AND QUICK SETUP 3. Bandpass filtering of test signal generators is absolutely necessary for SNR, THD and IMD tests. Note, a low noise signal generator along with a high Q bandpass filter is often necessary to achieve the attainable noise performance of the AD9260. 4. Test signal generators must have exceptional noise performance to achieve accurate SNR measurements. Good generators, together with fifth-order elliptical bandpass filters, are recommended for SNR tests. Narrow bandwidth crystal filters can also be used to filter generator broadband noise, but they should be carefully tested for operation at highsignal levels. 5. The analog inputs of the AD9260 should be terminated directly at the input pin sockets with the correct filter terminating impedance (50 or 75 ), or it should be driven by a low output impedance buffer. Short leads are necessary to prevent digital noise pickup. 6. A low noise (jitter) clock signal generator is required for good ADC dynamic performance. A poor generator can seriously impair good SNR performance particularly at higher input frequencies. A high-frequency generator, based on a clock source (e.g., crystal source), is recommended. Frequency-synthesized clock generators should generally be avoided because they typically provide poor jitter performance. See Note 8 if a crystal-based clock generator is used during FFT testing. A low jitter clock may be generated by using a high-frequency clock source and dividing this frequency down with a low noise clock divider to obtain the AD9260 input CLK. Maintaining a large amplitude clock signal may also be very beneficial in minimizing the effects of noise in the digital gates of the clock generation circuitry. Finally, special care should be taken to avoid coupling noise into any digital gates preceding the AD9260 CLK pin. Short leads are necessary to preserve fast rise times and careful decoupling should be used with these digital gates and the supplies for these digital gates should be connected to the same supplies as that of the internal AD9260 clock circuitry (Pins 44 and 38). 7. Two-tone testing will require isolation between test signal generators to prevent IMD generation in the test generator output circuits. 8. A very low side-lobe window must be used for FFT calculations if generators cannot be phase-locked and set to exact frequencies. 9. A well designed, clean PC board layout will assure proper operation and clean spectral response. Proper grounding and bypassing, short lead lengths, separation of analog and digital signals, and the use of ground planes are particularly important for high-frequency circuits. Multilayer PC boards are recommended for best performance, but if carefully designed, a two-sided PC board with large heavy (20 oz. foil) ground planes can give excellent results. 10. Prototype "plug-boards" or wire-wrap boards will not be satisfactory. * The AD9260 Evaluation Board is configured as follows when shipped: 1. 2.5 V external reference/4.0 V differential full-scale input: JP5, JP9 and JP10 connected, JP6 and JP7 disconnected. 2. 8x Mode/OSR: JP1 connected, JP2, JP3, and JP4 disconnected. 3. Full Speed Power Bias: R2 = 2 k and connected. 4. CSB pulled low: R6 = 49.9 and connected, R29 disconnected. 5. RESETB pulled high: R7 = 10 k and connected, R30 disconnected. 6. READ pulled high: R28 = 10 k and connected, R5 disconnected. 7. Single Tone Input: JP8 removed, input applied via IN-1 (J7). 8. Input signal common-mode level set by Trimpot R35 to 2.0 V: Jumper JP12 is disconnected and resistors Rx4 and Rx3 are connected. 9. AC Coupled Clock: JP12 connected and JP11 disconnected. Note: 50 terminated by R27. QUICK SETUP 1. Connect the required power supplies to the Evaluation Board as illustrated in Figure 22: 5 VA supplies to P5--Analog Power +5 VA supply to P4--Analog Power +5 VD supply to P3--Digital Power +5 VD supply to P2--Driver Power 2. Connect a Clock Source to CLKIN (J1): Note: 50 terminated by R1. 3. Connect an Input Signal Source to the IN-1 (J7). 4. Turn On Power! 5. The AD9260 Evaluation Board is now ready for use. APPLICATION TIPS 1. The ADC analog input should not be overdriven. Using a signal amplitude slightly lower than FSR will allow a small amount of "headroom" so that noise or DC offset voltage will not overrange the ADC and "hard limit" on signal peaks. 2. Two-tone tests can produce signal envelopes that exceed FSR. Set each test signal to slightly less than -6 dB to prevent "hard limiting" on peaks. REV. B -31- AD9260 RESET J5 CS J4 READ J3 DAV J2 DATA OUTPUT CONTROL BLOCK DRVDD U4 R7 10k R5 49.9 1 G1 19 G2 DRVDD 20 VCC 10 GND REFERENCE CONFIGURATION BLOCK JP9:EXT REF MDAVDD JP7:2.5V REF JP6:1V REF RD JP5:EXT REF R30 49.9 R29 10k R6 49.9 R28 10k JP15 MODE/OSR CONTROL BLOCK TP1:RD P1 33 P1 38 VREFEXT TP8:OTR JP4:1 74HC541 C11 0.1 F TP6 RESETB CSBBUE CT17 U2 TP3 33 32 31 30 29 28 27 26 25 24 23 REFCOM VREF SENSE RESET AVSS AVDD CS DAV OTR BIT01(MSB) BIT02 MDAVDD MDAVDD 2 3 4 5 6 7 8 9 A1 A2 A3 A4 A5 A6 A7 A8 Y1 Y2 Y3 Y4 Y5 Y6 Y7 Y8 18 17 16 15 14 13 12 11 JP3:4 C10 + 10 F CML JP2:2 1V JP1:8 TP2 C5 0.1 F R2 2k DRVDD DRVDD P1 39 P1 37 P1 35 C1 0.1 F C2 0.1 F C4 10 F TP4:REFB TP5:REFT 34 35 36 37 38 DVSS AVSS DVDD AVDD DRVSS DRVDD CLK READ BIT16(LSB) BIT15 BIT14 DVDD C8 0.1 F J1 CLKIN R27 49.9k R33 1k CT19 C61 10 F C62 0.1 F SHIELDED_TRACE CT20 DVDD FLAVDD RD Figure 71. Evaluation Board Top Level Schematic -32- AD9260 W1 VINA VINB W2 INVDD 1 2 3 4 5 6 7 8 9 10 11 C7 0.1 F R31 1k 39 40 41 42 43 44 CML C6 10 F JP11 DC COUPLED JP13 AC COUPLED C3 0.1 F 1 2 3 4 5 6 7 8 9 10 DIR A1 A2 A3 A4 A5 A6 A7 A8 GND VCC OUT_EN B1 B2 B3 B4 B5 B6 B7 B8 74HC245 20 19 18 17 16 15 14 13 12 11 CT1 CT2 CT3 CT4 CT5 CT6 CT7 CT8 P1 P1 P1 P1 P1 P1 P1 P1 1 3 5 7 9 11 13 15 TP7 TP9 TP11 TP12 TP13 TP15 MODE BIAS CAPB CAPT AVSS CML NC VINA VINB NC AVDD BIT03 BIT04 BIT05 BIT06 BIT07 BIT08 BIT09 BIT10 BIT11 BIT12 BIT13 22 21 20 19 18 17 16 15 14 13 12 CT18 DRVDD U3 DRVDD VCC OUT_EN B1 B2 B3 B4 B5 B6 B7 B8 74HC245 CT9 CT10 CT11 CT12 CT13 CT14 CT15 CT16 1 2 3 4 5 6 7 8 9 10 DIR A1 A2 A3 A4 A5 A6 A7 A8 GND 20 19 18 17 16 15 14 13 12 11 AGND P1 P1 P1 P1 P1 P1 P1 P1 P1 P1 P1 P1 P1 P1 P1 P1 P1 P1 P1 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 32 34 38 40 TP10 P1 P1 P1 P1 P1 P1 P1 P1 17 19 21 23 25 27 29 31 DATA OUTPUT BLOCK NC = NO CONNECT REV. B AD9260 R18 390 C20 0.1 F C9 TBD R19 390 J6 IN-2 R1 57.6 J7 IN-1 R15 57.6 R14 50 R16 390 2 3 8 VCC2 R23 390 VCC2 8 3 AD9632 R24 390 R32 390 VCC2 R34 390 IKPOT R35 1k RX3 XXX 3 C25 0.1 F R25 390 CX4 XXX U10 6 RX4 XXX VEE R26 390 2 5 4 7 U9 6 C24 100pF R21 390 VCC2 JP8 R22 390 VEE R17 390 5 4 AD9632 7 U7 8 3 AD9632 2 5 4 7 U8 6 C16 100pF VEE R20 390 6 JP16 R46 50 R47 50 VINA JP17 R48 50 C26 100pF R49 50 VINB 2 7 AD817R 4 JP12 9260CML + C23 10 F C22 0.1 F Figure 72. Evaluation Board Input Configuration Block P4:+5V 1 + C36 10 F L3 R40 C37 0.1 F R41 FLAVDD C38 0.1 F C39 0.01 F P5:+5AUX 1 + C27 47 F C55 0.1 F L6 R50 R51 VCC2 P5 L4 R42 C41 0.1 F R43 MDAVDD P5:-5AUX 2 1 + C56 47 F L7 R52 C57 0.1 F L2 R36 C29 0.1 F R53 VEE + C40 10 F C42 0.1 F C43 0.01 F P5 2 L5 R44 C45 0.1 F R45 INVDD P2:VDD 1 + C28 47 F R37 DRVDD + C44 10 F C46 0.1 F C47 0.01 F P2 2 P4 2 P3:D5 1 + C32 22 F L2 R38 C33 0.1 F R39 DVDD C34 0.1 F C35 0.01 F VEE C30 0.1 F VEE C31 0.1 F VEE C64 0.1 F VCC2 C48 0.1 F VCC2 C49 0.1 F VCC2 C50 0.1 F P3 2 U7 VCC2 U8 DRVDD U9 DRVDD U7 DRVDD C54 0.1 F U8 U9 EVALUATION BOARD POWER SUPPLY CONFIGURATION C51 0.1 F C52 0.1 F C53 0.1 F U10 U2 U3 U4 DEVICE SUPPLY DECOUPLING Figure 73. Evaluation Board Power Supply Configuration and Coupling REV. B -33- AD9260 Figure 74. Evaluation Board Component Side Layout (Not to Scale) Figure 75. Evaluation Board Solder Side Layout (Not to Scale) -34- REV. B AD9260 Figure 76. Evaluation Board Ground Plane Layout (Not to Scale) Figure 77. Evaluation Board Power Plane Layout (Not to Scale) REV. B -35- AD9260 OUTLINE DIMENSIONS Dimensions shown in millimeters and (inches). 44-Lead MQFP (S-44) C3197a-0-5/00 (rev. B) 00581 13.45 (0.529) 12.95 (0.510) 10.1 (0.398) 9.90 (0.390) 0 MIN 1 44 34 33 2.45 (0.096) MAX 1.03 (0.041) 0.73 (0.029) SEATING PLANE TOP VIEW (PINS DOWN) 8.45 (0.333) 8.3 (0.327) 0.25 (0.01) MIN 0.23 (0.009) 0.13 (0.005) 2.1 (0.083) 1.95 (0.077) 11 12 22 23 0.8 (0.031) BSC 0.45 (0.018) 0.3 (0.012) -36- REV. B PRINTED IN U.S.A. |
Price & Availability of AD9260
 |
|
|
|
|
All Rights Reserved © IC-ON-LINE 2003 - 2022 |
| [Add Bookmark] [Contact Us] [Link exchange] [Privacy policy] |
|
Mirror Sites : [www.datasheet.hk]
[www.maxim4u.com] [www.ic-on-line.cn]
[www.ic-on-line.com] [www.ic-on-line.net]
[www.alldatasheet.com.cn]
[www.gdcy.com]
[www.gdcy.net] |